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LTC3410-1.875 2.25MHz, 300mA Synchronous Step-Down Regulator in SC70
FEATURES

DESCRIPTIO
High Efficiency: Up to 93% Very Low Quiescent Current: Only 26A Low Output Voltage Ripple 300mA Output Current at VIN = 3V 380mA Minimum Peak Switch Current 2.5V to 5.5V Input Voltage Range 2.25MHz Constant Frequency Operation No Schottky Diode Required Stable with Ceramic Capacitors Shutdown Mode Draws < 1A Supply Current 2% Output Voltage Accuracy Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Available in Low Profile SC70 Package
The LTC (R)3410-1.875 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation is only 26A, dropping to <1A in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3410-1.875 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. Switching frequency is internally set at 2.25MHz, allowing the use of small surface mount inductors and capacitors. The LTC3410-1.875 is specifically designed to work well with ceramic output capacitors, achieving very low output voltage ripple and a small PCB footprint. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. The LTC3410-1.875 is available in a tiny, low profile SC70 package.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 5994885, 6127815, 6304066, 6498466, 6580258, 6611131.
APPLICATIO S

Cellular Telephones Wireless and DSL Modems Digital Cameras MP3 Players Portable Instruments
TYPICAL APPLICATIO
VIN 2.7V TO 5.5V VIN RUN VOUT GND SW
Efficiency and Power Loss vs Output Current
100 90
VOUT 1.875V
4.7H CIN 4.7F CER LTC3410-1.875
COUT 4.7F CER
EFFICIENCY VIN
80 70
EFFICIENCY (%)
60 50 40 30
34101875 TA01
20 10
4.2V
0 0.1
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2.7V 1 4.2V 3.6V 3.6V 0.01 2.7V POWER LOSS VIN 0.001 0.1
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POWER LOSS (W)
1 10 100 OUTPUT CURRENT (mA)
0.0001 1000
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LTC3410-1.875
ABSOLUTE
(Note 1)
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RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW RUN 1 GND 2 SW 3 6 VOUT 5 GND 4 VIN
Input Supply Voltage .................................. - 0.3V to 6V RUN, VOUT Voltages................................... - 0.3V to VIN SW Voltage (DC) ......................... - 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 500mA N-Channel Switch Sink Current (DC) ................. 500mA Peak SW Sink and Source Current .................... 630mA Operating Temperature Range (Note 2) .. - 40C to 85C Junction Temperature (Notes 3, 5) ...................... 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
ORDER PART NUMBER LTC3410ESC6-1.875 SC6 PART MARKING LCFQ
SC6 PACKAGE 6-LEAD PLASTIC SC70
TJMAX = 125C, JA = 250C/ W
Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25C. VIN = 3.6V unless otherwise specified.
SYMBOL IVOUT IPK VOUT VOUT VLOADREG VIN VUVLO IS PARAMETER Output Voltage Feedback Current Peak Inductor Current Regulated Output Voltage Output Voltage Line Regulation Output Voltage Load Regulation Input Voltage Range Undervoltage Lockout Threshold Input DC Bias Current Burst Mode(R) Operation Shutdown Oscillator Frequency RDS(ON) of P-Channel FET RDS(ON) of N-Channel FET SW Leakage RUN Threshold RUN Leakage Current VIN Rising VIN Falling (Note 4) VOUT = 1.945V, ILOAD = 0A VRUN = 0V VOUT = 1.875V VOUT = 0V ISW = 100mA ISW = -100mA VRUN = 0V, VSW = 0V or 5V, VIN = 5V

CONDITIONS
MIN 380
TYP 3.3 500 1.875 0.04 0.5
MAX 6 1.913 0.4 5.5
UNITS A mA V %/V % V V V A A MHz kHz A V A
VIN = 3V, VOUT = 1.64V, Duty Cycle < 35% VIN = 2.5V to 5.5V ILOAD = 50mA to 250mA

1.837
2.5 2 1.94 26 0.1 1.8 2.25 310 0.75 0.55 0.01 0.3 1 0.01
2.3
35 1 2.7 0.9 0.7 1 1.5 1
fOSC RPFET RNFET ILSW VRUN IRUN
Burst Mode is a registered trademark of Linear Technology Corporation. Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3410E-1.875 is guaranteed to meet performance specifications from 0C to 70C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3410-1.875: TJ = TA + (PD)(250C/W) Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.
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LTC3410-1.875 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1) Efficiency vs Input Voltage
100 90 80 IOUT = 250mA 70 60 50 40 30 2.5 IOUT = 0.1mA IOUT = 100mA
EFFFICIENCY (%)
IOUT = 1mA
EFFICIENCY (%)
3
4.5 4 3.5 INPUT VOLTAGE (V)
Output Voltage vs Temperature
1.911 VIN = 3.6V
2.7 2.6
OSCILLATOR FREQUENCY (MHz)
1.899
OUTPUT VOLTAGE (V)
1.887 1.875 1.863 1.851 1.839 -50 -25
50 25 75 0 TEMPERATURE (C)
Oscillator Frequency vs Supply Voltage
2.7 2.6
OSCILLATOR FREQUENCY (MHz)
OUTPUT VOLTAGE (V)
2.5 2.4 2.3 2.2 2.1 2.0 1.9 1.8 2 3 5 4 SUPPLY VOLTAGE (V) 6
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UW
Efficiency vs Output Current
100 90 80
IOUT = 10mA
70 60 50 40 30 20 10 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 10 100 OUTPUT CURRENT (mA) 1000
34101875 G02
5
5.5
0 0.1
34101875 G01
Oscillator Frequency vs Temperature
VIN = 3.6V
2.5 2.4 2.3 2.2 2.1 2.0 1.9
100
125
1.8 -50
-25
0 25 50 75 TEMPERATURE (C)
100
125
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Output Voltage vs Load Current
1.900 1.895 1.890 1.885 1.880 1.875 1.870 1.865 1.860 1.855 1.850 0 200 100 300 LOAD CURRENT (mA) 400
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LTC3410-1.875 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1) RDS(ON) vs Input Voltage
1.2 1.1 1.0 0.9
RDS (ON) ()
0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 1 2
RDS (ON) ()
SYNCHRONOUS SWITCH
5 4 3 INPUT VOLTAGE (V)
Dynamic Supply Current vs VIN
50 ILOAD = 0A
DYNAMIC SUPPLY CURRENT (A)
40
DYNAMIC SUPPLY CURRENT (A)
30
20
10
0
1
2
Switch Leakage vs Temperature
110 100 90
SWITCH LEAKAGE (nA)
VIN = 5.5V RUN = 0V
80 70 60 50 40 30 20 10 0 -50 -25 50 25 0 75 TEMPERATURE (C) 100 125 MAIN SWITCH SYNCHRONOUS SWITCH
LEAKAGE CURRENT (pA)
4
UW
3 VIN (V)
RDS(ON) vs Temperature
1.2 1.0 VIN = 4.2V VIN = 2.7V VIN = 3.6V
MAIN SWITCH
0.8 0.6 VIN = 4.2V 0.4 VIN = 2.7V 0.2 VIN = 3.6V MAIN SWITCH SYNCHRONOUS SWITCH
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7
0 -50 -30 -10 10 30 50 70 90 110 130 TEMPERATURE (C)
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Dynamic Supply Current vs Temperature
50
40
30
20
10
4
5
6
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0 -50 -25
50 25 0 75 TEMPERATURE (C)
100
125
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Switch Leakage vs Input Voltage
600 550 500 450 400 350 300 250 200 150 100 50 0 0 1 4 3 2 INPUT VOLTAGE (V) 5 6 SYNCHRONOUS SWITCH MAIN SWITCH
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LTC3410-1.875 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1) Burst Mode Operation Start-Up from Shutdown
SW 5V/DIV VOUT 50mV/DIV AC COUPLED IL 100mA/DIV 2s/DIV
VIN = 3.6V ILOAD = 10mA
Start-Up from Shutdown
RUN 2V/DIV VOUT 1V/DIV IL 200mA/DIV VIN = 3.6V ILOAD = 0A 200s/DIV
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RUN 2V/DIV VOUT 1V/DIV IL 200mA/DIV 200s/DIV
34101875 G13
VIN = 3.6V ILOAD = 300mA
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Load Step
VOUT 100mV/DIV AC-COUPLED IL 200mA/DIV ILOAD 200mA/DIV 10s/DIV VIN = 3.6V ILOAD = 0mA TO 300mA
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Load Step
VOUT 100mV/DIV AC-COUPLED IL 200mA/DIV ILOAD 200mA/DIV 10s/DIV VIN = 3.6V ILOAD = 15mA TO 300mA
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LTC3410-1.875
PI FU CTIO S
RUN (Pin 1): Run Control Input. Forcing this pin above 1.5V enables the part. Forcing this pin below 0.3V shuts down the device. In shutdown, all functions are disabled drawing <1A supply current. Do not leave RUN floating. GND (Pins 2, 5): Ground Pin. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2F or greater ceramic capacitor. VOUT (Pin 6): Output Voltage Feedback. An internal resistive divider divides the output voltage down for comparison to the internal 0.8V reference voltage.
FU CTIO AL DIAGRA
SLOPE COMP OSC OSC
VOUT 6
FREQ SHIFT R1 322.5k R2 240k
0.8V VFB
EA S R
VIN RUN 1 0.8V REF
SHUTDOWN
6
-
IRCMP
+
W
-
+
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0.65V
4 VIN
- +
0.4V
- +
EN SLEEP
-
BURST Q Q SWITCHING LOGIC AND BLANKING CIRCUIT
ICOMP
+
5
RS LATCH
ANTISHOOTTHRU
3 SW
5 2 GND
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LTC3410-1.875
OPERATIO
Main Control Loop The LTC3410-1.875 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. The VOUT pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from the internal resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which in turn, causes the EA amplifier's output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle. Burst Mode Operation The LTC3410-1.875 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 70mA regardless of the output load. Each burst event can last from a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 26A. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage droops, the EA amplifier's output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET on. This process repeats at a rate that is dependent on the load demand.
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(Refer to Functional Diagram)
Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 310kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator's frequency will progressively increase to 2.25MHz when VOUT rises above 0V. Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3410-1.875 uses a patented scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles.
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LTC3410-1.875
APPLICATIO S I FOR ATIO
The basic LTC3410-1.875 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT.
VIN 2.7V TO 5.5V 4.7H CIN 4.7F CER VIN RUN VOUT GND
34101875 F01
SW COUT 4.7F CER
LTC3410-1.875
Figure 1. High Efficiency Step-Down Converter
Inductor Selection For most applications, the value of the inductor will fall in the range of 2.2H to 4.7H. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is IL = 120mA (40% of 300mA). IL = V 1 VOUT 1- OUT ( f)(L) VIN (1)
The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 360mA rated inductor should be enough for most applications (300mA + 60mA). For better efficiency, choose a low DC-resistance inductor. The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 100mA. Lower inductor values (higher IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase.
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Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3410-1.875 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3410-1.875 applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER Taiyo Yuden CB2016T2R2M CB2012T2R2M LBC2016T3R3M ELT5KT4R7M CDRH2D18/LD NR30102R2M NR30104R7M FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D MAX DC VALUE CURRENT DCR HEIGHT 2.2H 2.2H 3.3H 4.7H 4.7H 2.2H 4.7H 4.7H 3.3H 2.2H 510mA 530mA 410mA 950mA 450mA 0.13 1.6mm 0.33 1.25mm 0.27 1.6mm 0.2 1.2mm 0.2 2mm
VOUT 1.875V
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Panasonic Sumida Murata Taiyo Yuden FDK
630mA 0.086 2mm 1100mA 0.1 1mm 750mA 0.19 1mm 1100mA 0.11 1mm 1200mA 0.1 1mm 1300mA 0.08 1mm
LQH32CN4R7M23 4.7H
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LTC3410-1.875
APPLICATIO S I FOR ATIO
CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
CIN required IRMS IOMAX
[VOUT (VIN - VOUT )]1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer's ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple VOUT is determined by:
1 VOUT IL ESR + 8fC OUT
where f = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since IL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations.
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Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC34101.875's control loop does not depend on the output capacitor's ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. The recommended capacitance value to use is 4.7F for both the input and output capacitors.
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LTC3410-1.875
APPLICATIO S I FOR ATIO
Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3410-1.875 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 2. 1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is
1 VIN = 3.6V 0.1
POWER LOSS (W)
0.01
0.001
0.0001
0.00001 0.1
Figure 2. Power Loss vs Load Current
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switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is "chopped" between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 - DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss.
10 100 1 LOAD CURRENT (mA) 1000
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LTC3410-1.875
APPLICATIO S I FOR ATIO
Thermal Considerations In most applications the LTC3410-1.875 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3410-1.875 is running at high ambient temperature with low supply voltage, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150C, both power switches will be turned off and the SW node will become high impedance. To prevent the LTC3410-1.875 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(JA) where PD is the power dissipated by the regulator and JAis the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3410-1.875 with an input voltage of 2.7V, a load current of 300mA and an ambient temperature of 70C. From the typical performance graph of switch resistance, the R DS(ON) of the P-channel switch at 70C is approximately 1.05 and the RDS(ON) of the N-channel synchronous switch is approximately 0.75. The series resistance looking into the SW pin is: RSW = 1.05 (0.69) + 0.75 (0.31) = 0.96 Therefore, power dissipated by the part is: PD = ILOAD2 * RDS(ON) = 86.4mW
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For the SC70 package, the JA is 250C/ W. Thus, the junction temperature of the regulator is: TJ = 70C + (0.0864)(250) = 91.6C which is well below the maximum junction temperature of 125C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (ILOAD * ESR), where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1F) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 * CLOAD). Thus, a 10F capacitor charging to 3.3V would require a 250s rise time, limiting the charging current to about 130mA.
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LTC3410-1.875
APPLICATIO S I FOR ATIO
PC Board Layout Checklist
When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3410-1.875. These items are also illustrated graphically in Figures 3 and 4. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 3. Keep the (-) plates of CIN and COUT as close as possible. Design Example As a design example, assume the LTC3410-1.875 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.3A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low
1
RUN LTC3410-1.875 6
2
-
VOUT COUT
GND
VOUT VIN 5 CIN
+
3 L1
SW
4 SW VIN COUT
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BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 3. LTC3410-1.875 Layout Diagram
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and high load currents is important. With this information we can calculate L using Equation (1), L= V 1 VOUT 1- OUT ( f)(IL ) VIN (3) Substituting VOUT = 1.875V, VIN = 4.2V, IL = 100mA and f = 2.25MHz in Equation (3) gives:
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L=
1 . 875V 1 . 875V 1- = 4 .6H 6 2 . 25MHz(100mA) 4 . 2V
A 4.7H inductor works well for this application. For best efficiency choose a 360mA or greater inductor with less than 0.3 series resistance. CIN will require an RMS current rating of at least 0.125A ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.5. In most cases, a ceramic capacitor will satisfy this requirement. Figure 5 shows the complete circuit along with its efficiency curve.
VOUT
VIA TO VIN
VIN
PIN 1 L1 LTC34101.875
CIN
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Figure 4. LTC3410-1.875 Suggested Layout
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LTC3410-1.875
APPLICATIO S I FOR ATIO
VIN 2.7V TO 4.2V
CIN 4.7F CER
TAIYO YUDEN JMK212BJ475 *MURATA LQH32CN4R7M23
100 90 80 70
EFFICIENCY (%)
60 50 40 30 20 10 0 0.1 EFFICIENCY, VIN = 2.7V EFFICIENCY, VIN = 3.6V EFFICIENCY, VIN = 4.2V 1 10 LOAD (mA) 100 1000
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VOUT 100mV/DIV AC COUPLED IL 200mA/DIV ILOAD 200mA/DIV
20s/DIV VIN = 3.6V ILOAD = 100mA TO 300mA
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4 VIN RUN VOUT GND 2, 5
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SW
3
4.7H* COUT 4.7F CER
VOUT 1.875V
LTC3410-1.875 1 6
Figure 5a
Figure 5b
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Figure 5c
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LTC3410-1.875
TYPICAL APPLICATIO
VIN 2.7V TO 4.2V
TAIYO YUDEN JMK212BJ475 *FDK MIPF2520D
EFFICIENCY (%)
14
U
Using Low Profile Components, <1mm Height
4.7H* COUT 4.7F CER 4 CIN 4.7F VIN RUN VOUT GND 2, 5
34101875 TA03
SW
3
VOUT 1.875V
LTC3410-1.875 1 6
Low Profile Efficiency
100 VIN = 2.7V VIN = 3.6V VIN = 4.2V
90
80
70
60
50 0.1
1
10 LOAD (mA)
100
1000
34101875 TA04
Load Step
VOUT 100mV/DIV AC COUPLED IL 200mA/DIV ILOAD 200mA/DIV
20s/DIV VIN = 3.6V ILOAD = 100mA TO 300mA
34101875 TA05
34101875f
LTC3410-1.875
PACKAGE DESCRIPTIO
0.47 MAX
0.65 REF
2.8 BSC 1.8 REF
RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.10 - 0.40
GAUGE PLANE 0.15 BSC 0.26 - 0.46
NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. DETAILS OF THE PIN 1 INDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE INDEX AREA 7. EIAJ PACKAGE REFERENCE IS EIAJ SC-70 8. JEDEC PACKAGE REFERENCE IS MO-203 VARIATION AB
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
U
SC6 Package 6-Lead Plastic SC70
(Reference LTC DWG # 05-08-1638)
1.80 - 2.20 (NOTE 4) 1.00 REF 1.80 - 2.40 1.15 - 1.35 (NOTE 4) INDEX AREA (NOTE 6) PIN 1 0.65 BSC 0.15 - 0.30 6 PLCS (NOTE 3) 0.80 - 1.00 0.00 - 0.10 REF 1.00 MAX 0.10 - 0.18 (NOTE 3)
SC6 SC70 1205 REV B
34101875f
15
LTC3410-1.875
RELATED PARTS
PART NUMBER LT1616 LT1676 LTC1701/LTC1701B LT1776 LTC1877 LTC1878 LTC1879 LTC3403 LTC3404 LTC3405/LTC3405A LTC3406/LTC3406B LTC3409 LTC3410/LTC3410B LTC3411 LTC3412 LTC3440 DESCRIPTION 500mA (IOUT), 1.4MHz, High Efficiency Step-Down DC/DC Converter 450mA (IOUT), 100kHz, High Efficiency Step-Down DC/DC Converter 750mA (IOUT), 1MHz, High Efficiency Step-Down DC/DC Converter 500mA (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter with Bypass Transistor 600mA (IOUT), 1.4MHz, Synchronous Step-Down DC/DC Converter 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 1.5MHz/2.25MHz, Synchronous Step-Down DC/DC Converter 300mA (IOUT), 2.25MHz, Synchronous Step-Down DC/DC Converter 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter COMMENTS 90% Efficiency, VIN = 3.6V to 25V, VOUT = 1.25V, IQ = 1.9mA, ISD = <1A, ThinSOT Package 90% Efficiency, VIN = 7.4V to 60V, VOUT = 1.24V, IQ = 3.2mA, ISD = 2.5A, S8 Package 90% Efficiency, VIN = 2.5V to 5V, VOUT = 1.25V, IQ = 135A, ISD = 1A, ThinSOT Package 90% Efficiency, VIN = 7.4V to 40V, VOUT = 1.24V, IQ = 3.2mA, ISD = 30A, N8, S8 Packages 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 10A, ISD = <1A, MS8 Package 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10A, ISD = <1A, MS8 Package 95% Efficiency, VIN = 2.7V to 10V, VOUT = 0.8V, IQ = 15A, ISD = <1A, TSSOP-16 Package 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = Dynamically Adjustable, IQ = 20A, ISD = <1A, DFN Package 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 10A, ISD = <1A, MS8 Package 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 20A, ISD = <1A, ThinSOT Package 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.6V, IQ = 20A, ISD = <1A, ThinSOT Package 95% Efficiency, VIN = 1.6V to 5.5V, VOUT = 0.613V, IQ = 65A, DD8 Package 96% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 26A, SC70 Package 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60A, ISD = <1A, MS Package 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60A, ISD = <1A, TSSOP-16E Package 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 2.5V, IQ = 25A, ISD = <1A, MS Package
34101875f
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
LT 0306 * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2005

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