View AN993_912376.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

AN993 APPLICATION NOTE
ELECTRONIC BALLAST WITH PFC
USING L6574 AND L6561
by F. Sandrini & U. Moriconi
The advent of dedicate IC for lamp ballast applications is replacing the old solutions based on bipolar transistor driven by a saturable pulse transformer. The L6574 is a high performance ballast driver, designed in 600V BCD OFF-LINE technology, which ensures all the features needed to drive and control properly a fluorescent bulb. It is provided with a builtin VCO start up sequence circuitry, Protections, and an OP-AMP, useful to implement a closed loop control of the lamp current.
INTRODUCTION Half bridge converter for electronic lamp ballast Voltage fed series resonant half bridge inverters are currently used for fluorescent lamps (fig.1). This topology allows to easily operate in Zero Voltage Switching (ZVS) resonant mode, drammatically reducing the transistor switching losses and the electromagnetic interference. To design a cost effective, compact and smart electronic lamp ballast it could be used a dedicated IC able to drive directly the power MOSFETs of the half bridge. Such controllers require a high voltage capability for the high side floating transistor driver.
Figure 1. Half Bridge topology
HV
DRIVER
LRES
CRES
D98IN819
Lamp requirements To provide long life and to insure an efficient ignition of the lamp the cathodes must be preheated. In fact the preheating of the filaments allows an easy strike of the lamp reducing ignition voltage. During preheating time the lamp is characterised by a high impedance and the current flows only in the filaments. The resistance value of the filaments are strictly dependent on the lamp model. Typically these filaments present a initial low value
May 2002
1/20
AN993 APPLICATION NOTE
(a few Ohms) that will increase by 4-5 times during the preheating. After the preheating time the lamp must be ignited, by increasing the voltage across it. The ignition voltage value also depends on the lamp type, and it increases with the aging. For a typical TL 58W it is not much less than 1000V. Using a simple inverter, with a constant switching frequency, external circuitry must be used (e.g. PTC or discrete timer). Instead with the ST L6574 smart controller both the preheating and the ignition functions are achieved by using simple resistors and a capacitor, which set all the start-up procedure. L6574 Ballast Driver The L6574, whose internal block diagram is shown in fig.2, is an IC intended to drive two power MOS or IGBT in half bridge topology, ensuring all the features needed to drive and control properly a fluorescent bulb. Moreover, by varying the switching frequency it is possible to modulate the current in the lamp, therefore the output power. The device is available in DIP16 and SO16N packages. The most significant features of the L6574 concern the following points:
s s s s s s s s s s s s
high voltage rail up to 600V; dV/dt immunity 50 V/ns in full temperature range; driver current capability (250 mA source and 450 mA sink); switching times 80/40 ns rise fall with 1 nF load; CMOS shut down input; under voltage lock out; preheat and frequency shifting timing; sense OP AMP for closed loop control or protection features; high accuracy current controlled oscillator; integrated bootstrap diode; clamping on VS; SO16, DIP16 package
Figure 2. Internal Block diagram of the L6574
H.V.Bus
VS
OPOUT OPINOPIN+
5 6
OP AMP + -
12
16 Vboot BOOTSTRAP DRIVER 15 HVG
UV DETECTION
HVG DRIVER
Cboot LOAD
7 Imin 4 VREF DEAD TIME Ifs Imax 2 CONTROL LOGIC 3 VCO Cf 1 Cpre VREF + Rpre + Vthpre + + VTHE VTHE Ipre LVG DRIVER LEVEL SHIFTER VS
14 OUT DR IVING LOGIC
Ri gn
11
LVG
10 GND
8
EN1
9
EN2
2/20
AN993 APPLICATION NOTE
Figure 3. Typical Application's Connection
+ HVBus
D1
CSnub
CSupply D2
CBoot Qh
12 16 15 Rgh L_ballast 7 14
2 R1
Rpre.
6 11
Ql Cb
L6574
8 9 10
Rgl R5
5 D3 Ref. 43 R2 1
R6 D4
LAMP
Cres
C3 C1 C2 R3 Rcs C4
Rign.
Cf
Cpre
R4
R7
R9
R8
Device Pins Description
N. 1 Name CPRE Function Preheat Timing Capacitor. The capacitor CPRE sets the preheating and the frequency shift time, according to the relations: tPRE = KPRE * C PRE and tSH = KFS * CPRE (typ. KPRE = 1.5s/ F, K FS = 0.15s/F). This feature is obtained by charging CPRE with two different currents. During tPRE this current is independent of the external components, so CPRE is charged up to 3.5V (preheat timing comparator threshold). During tSH the current depends on RPRE value (i.e. on the difference between fPRE and fIGN). In this way tSH is always set at 0.1t PRE. In steady state the voltage at pin 1 is 5V (see fig.5). Maximum Oscillation Frequency Setting. The resistance connected between this pin and ground sets the fPRE value, fixing the difference between fPRE and fIGN (fPRE > fIGN). At the end of the Start-up procedure, the effect current drown from RPRE is over. The voltage at this pin is fixed at VREF =2V. Oscillator Frequency Setting. The capacitor CF, along with to RPRE and RIGN, sets fPRE and fING. In normal operation this pin shows a triangular wave. Minimum Oscillation Frequency Setting. The resistance connected between this pin and ground sets the fIGN value. The voltage at this pin is fixed at VREF =2V.
2
RPRE
3
CF
4
RIGN
3/20
AN993 APPLICATION NOTE
Device Pins Description (continued)
N. 5 6 7 8 Name OPout OPinOPin+ EN1 Function Out of the operational amplifier. To implement a feedback control loop this pin can be connected to the RIGN pin by means an appropriate circuitry. Inverting Input of the operational amplifier. Non Inverting Input of the operational amplifier. Enable 1. This pin (active high), forces the device in a latched shutdown state (like in the under voltage conditions). There are two ways to resume normal operation: - the first is to reduce the supply voltage below the undervoltage threshold and then increase it again until the valid supply is recognised. - the second is activating EN2 input (see fig. 10). The enable 1 is especially designed for strong fault (e.g. in case of lamp disconnection). Enable 2. EN2 input (active high) restarts the start-up procedure (preheating and ignition sequence). This features is useful if the lamp does not turn-on after the first ignition sequence (see fig. 9). Ground. Low Side Driver Output. This pin must be connected to the low side power MOSFET gate of the half bridge. A resistor connected between this pin and the power MOS gate can be used to reduce the peak current. Supply Voltage. This pin, connected to the supply filter capacitor, is internally clamped (15.6V typical). Non Connected. This pin set a distance between the pins related to the HV and those related to the LV side. High Side Driver Floating Reference. This pin must be connected close to the source of the high side power MOS or IGBT. High Side Driver Output. This pin must be connected to the high side power MOSFET gate of the half bridge. A resistor connected between this pin and the power MOS gate can be used to reduce the peak current. Bootstrapped Supply Voltage. Between this pin and VS must be connected the bootstrap capacitor. A patented integrated circuitry replaces the external bootstrap diode, by means of a high voltage DMOS, synchronously driven with the low side power MOSFET.
9
EN2
10 11
GND LVG
12 13 14 15
VS N.C. OUT HVG
16
VBOOT
4/20
AN993 APPLICATION NOTE
DEVICE BLOCK DESCRIPTION The preheating control section and the bootstrap section are tightly connected to application design.Here below there are some details on their working and how they have to be used. Preheating and ignition section L6574 turn on sequence is divided into three phases: preheating, ignition and normal operation (fig. 4). Preheating phase is characterised by the highest oscillation frequency (fmax) for a period Tpre. During ignition phase the frequency shifts from fmax to fmin (that is the normal operation frequency) in a period Tsh.
Figure 4. Start-Up Timing Diagram & EN2 Function
Power-O.K.
VSupply
V(Cpre)
preheating
Osc. freq.
steady state operation ignition
fMAX
fMIN
IL
TPRE = K * CPRE TIGN = 0.1 * TPRE
TPRE
Time
All the parameters specified above are set choosing properly few external components. Tpre and Tsh are set by means of the capacitor Cpre that is connected at pin 1. During the preheating time, Tpre, the capacitor Cpre is charged by means of a constant current Ipre internally generated, which doesn't depend on the external components. The voltage across Cpre increases linearly up to the "preheating threshold", in which the preheating phase finishes.
V th T pre = C pre -------- , where Vth = 3.5V and Ipre = 2.3A I pre
That is to say: Tpre = 1.5s/F * Cpre
5/20
AN993 APPLICATION NOTE
Figure 5. Timing block
Ifs
Ipre
Cpre
DISCHARGE
Figure 6. Timing Oscillator block
TIMING
Imax
Imin
gm Cpre
Iosc
OSC
After the preheating time the capacitor Cpre is first quickly discharged, and then recharged by the current Ifs, generating a second voltage ramp which fed a transconductance amplifier, as shown in fig. 6 (the switch is closed). Thus this voltage signal is converted in a growing current which is subtracted to Imax, to produce the frequency shifting from fmax to fmin. The current which drives the oscillator to set the frequency during the shifting is equal to:
gm Ifs I o sc = I min + ( Im ax - gm V Cpre ( t - Tp re ) ) = I m in + I ma x - ------------- ( t - T p re ) [a] C pre
Where:
V REF VR EF --------I m in = ---- -, Im ax = --------------, VR EF = 2V R ign R pre
Rign and Rpre are the resistors connected to pin 4 and pin 2. At the end of preheating time (t = Tpre) L6574 oscillates at fmax, set by:
1 1 I osc ( T pre ) = I osc 0 ) = Im in = Im ax = V REF ---------- + ----------- R R pre ig n
That means that the preheating frequency depends on both Rpre and Rign. At the end of the frequency shifting (t = Tpre + Tsh), the second term of eq.[a] decreases to zero and the switching frequency is set only by Imin (i.e. Rign):
6/20
AN993 APPLICATION NOTE
V REF I osc ( T sh ) = Im in = ------------R ign
Since the second term of eq. [a] is equal to zero, we have:
g m I fs C pre Ima x -T Im ax - ------------- sh = 0 T sh = ----------------------- [b] Cp re gm I fs
Note that there is not a fixed threshold of the voltage across Cpre in which the ignition phase finishes (i.e. the end of the frequency shifting): Tsh depends on Cpre, Imax, gm, and Ifs (eq. [b]). This fact can be seen also in fig. 7. Making Tsh independent of Imax, the Ifs current has been designed to be a fraction of Imax, so:
I max C pre I max C pre Ifs = ----------- T sh = ------------------------ T sh = ----------- T sh = k fs C p re K g m Im ax K gm K
In this way the frequency shift time depends only by the capacitor Cpre. The typical value of the kfs constant (Frequency Shift Timing Constant) is 0.15 s/F, that is: kfs = kpre/10. So choosing Cpre both Tpre and Tsh are set. The frequencies fmin and fmax depend on the resistors Rpre and Rign, but also on the capacitor Cf (Oscillator frequency setting -> capacitor at pin 3). fmin is set choosing Cf and Rign, then with Rpre the f = fmax - fmin is set. Simplified equations can be used:
1.41 -----------------f min = ---Rig n C f 1.41 ( Rpre + Rign ) f m ax = -------------------------------------------------- R pre R ign C f
These equation fit well with measured values, especiallyin the frequencies range 30 to 100kHz: in fig. 8 there is a comparison between fmin measured and computed data (@ Cf = 470pF).
Figure 7. Cpre voltage and frequency shifting
V(Cpre)
Figure 8. Operating Frequency @ Cf = 470pF
120
measured values computed values
100
time freq preheating ignition
fmin(kHz)
80
60
40
time
20 40 60 80 100 120
Rign (kW)
7/20
AN993 APPLICATION NOTE
To summarise:
parameter Tpre Tsh f = fmax - fmin fmin set choosing C pre C pre R pre Rign & C f
If Rpre is not connected at pin 2, there is no f , so fmax and fmin have the same value. Moreover there is no Imax, that is to say there is no Ifs. In this case Cpre is charged only once by Ipre = 2.3A up to 3.5V (fig 8: there is only the first voltage raising) . When Cpre is discharged there is no current Ifs to charge it once more. If Rpre has been connected when preheating and ignition phases are ended, pin 1(Cpre) is at 4.8-5V with a current capability of few A (1/6 IRpre). If Rpre is not connected, Cpre is at GND level in normal operation phase.
Controls section. L6474 is provided with two control's functions, EN1 and EN2. Both the functions are active high. To well understand the effect of this functions, please refer to figures Fig. 9 and Fig. 10. - EN1 (latched enable) is dedicated to stop all the functions of the device and it is usually activated in case of strong fault (i.e. lamp disconnection). It is possible to cancel this status, in two different way: reducing the supply voltage below its minimum threshold or activating the second control pin EN2 (EN2 overrides EN1). As the device start again restarts to operate, the start-up sequence will be reactivated. - EN2 (Repeating start-up) is dedicated to restart the preheating and ignition procedure; it is normally activated when the application fails to ignite the lamp. As the EN2 is activated the starting procedure restarts. To be considered that the L6574 operates to the maximum frequency for all the time that EN2 is active.
Figure 9. Start-Up Timing Diagram & EN2 Function
VSupply
preheating
V(Cpre)
fMAX
ignition preheating fMIN Foult-ignition
ignition
EN2
IL
TPRE
TIGN(TSH)
Time
8/20
AN993 APPLICATION NOTE
Figure 10. Controls Timing Diagram
VSupply
LVG
HVG
Latched enable
EN1
EN2
Forced restart
Time
Bootstrap section A bootstrap circuitry is needed to supply the high voltage section. This function is normally accomplished by a high voltage fast recovery diode (fig. 11a). In the L6574 a patented integrated structure replaces the external diode. It is realised by a high voltage DMOS, driven synchronously with the low side driver (LVG), with in series a diode, as shown in fig. 11b An internal charge pump (fig. b) provides the DMOS driving voltage . The diode connected in series to the DMOS has been added to avoid undesirable turn on of it. Cboot selection and charging: To choose the proper Cboot value the external MOS can be seen as an equivalent capacitor. This capacitor Cext is related to the MOS total gate charge :
Q gate Cext = --------------Vgate
The ratio between the capacitors Cext and Cboot is proportional to the cyclical voltage loss . It has to be: Cboot>>>Cext e.g.: if Qgate is 30nC and Vgate is 10V, Cext is 3nF. With Cboot = 100nF the drop would be 300mV. If HVG has to be supplied for a long time, the Cboot selection has to take into account also the leakage losses. The internal bootstrap driver gives great advantages: the external fast recovery diode can be avoided (it usually has great leakage current). This structure can work only if Vout is close to GND (or lower) and in the meanwhile the LVG is on. The charging time (Tcharge) of the Cboot is the time in which both conditions are fulfilled and it has to be long enough to charge the capacitor. The bootstrap driver introduces a voltage drop due to the DMOS Rdson (typical value @25 is 150 Ohm). At low frequency this drop can be neglected. Anyway increasing the frequency it must be taken in to account. The following equation is useful to compute the drop on the bootstrap DMOS:
Qga te Vd rop = I ch arg e R d so n V drop = --------------------R dson T c h arg e
9/20
AN993 APPLICATION NOTE
where Qgate is the gate charge of the external power MOS, Rdson is the on resistance of the bootstrap DMOS, and Tcharge is the charging time of the bootstrap capacitor. For example: using a power MOS with a total gate charge of 30nC the drop on the bootstrap DMOS is about 1V, if the Tcharge is 5 s. In fact:
30nC Vd rop = -------------- 150 ~ 0.9V 5s
Vdrop has to be taken into account when the voltage drop on Cboot is calculated: if this drop is too high, or there is not a sufficient charging time, an external diode can be used.
Figure 11. Bootstrap driver
DBOOT
VS
VBOOT H.V. HVG
VS
VBOOT H.V. HVG
CBOOT VOUT TO LOAD
CBOOT VOUT TO LOAD
LVG
LVG
a
b
EX_D99IN1056
DEMO APPLICATION DESCRIPTION The design has been developed to drive a TL fluorescent lamp up to 58W. It is composed of two sections: the PFC, using the L6561 controller, and the ballast, based on the L6574 (see fig.9 and fig.10). The application is provided with a current feedback, that allows power control (and in case the dimming function) by varying the switching frequency during the normal lamp burning. The application is also provided with a safety circuitry, that acts in case of open load or faulty ignition of the lamp.
10/20
Fuse D1 BYT11600
Bridge
T1 1.24mH (E25*13*7) 10 Turns 1.3mm gapped
Figure 12. Demo Application circuit
C1 220nF 630V R16 47 D2 1N4148 C5 100nF C20 100nF R21 22 2 R17 9.1K 7 14 R22 22 6 R10 750K 4 1 R11 750K R14 4.7K 43 R18 100K C8 100nF R19 100K C13 470pF C14 1F R24 6.8K C15 330nF R25 0.68 RSENSE R8 0.68 R12 9.53K R9 0.68 R15 1.5K 1 D3 1N4148 C7 22F 450V 5 C9 8.2nF R13 100K 12 16 15 C6 4.7F 25V DZ1 14V C11 680pF 630V C12 100nF R20 47K
R1 1.5M
R5 68K
NTC
R3 120K
R4 120K
Q2 STP4NB50/ STP6NB50
5 8 R6 22 7 R7 47K
Q1 STP6NB50
6
L1 2.1mH C17 100nF 250V R23 47K R26 390K R28 750K C18 100nF 250V
C2 220nF 400V
L6561
11 8
3
2
Q3 STP4NB50/ STP6NB50
L6574
9 10
R29 750K D4 LAMP 1N4148
C19 8.2nF 1500V
C4 680nF
R2 10.1K
C3 10nF
R30 3.9K
R27 6.8K
R33 9.1K
D99IN1064
C16 1F
R32 20K
AN993 APPLICATION NOTE
11/20
AN993 APPLICATION NOTE
Figure 13. PCB and components layout.
37mm Component Layer Solder Layer
220mm
12/20
AN993 APPLICATION NOTE
Power factor section Even if the PFC stage is not strictly necessary for electronic ballast application, in this design it has been introduced for the following reasons. The front-end stage of conventional off-line converters, typically made up of a full wave rectifier bridge with a capacitor filter, gets an unregulated DC bus from the AC mains. Therefore the instantaneous line voltage is below the voltage on the capacitor most of the time, thus the rectifiers conduct only for a small portion of each line half-cycle. The current drawn from the mains is then a series of narrow pulses whose amplitude is 5-10 times higher than the resulting DC value. Lots of drawbacks result from that: much higher peak and RMS current drawn from the line, distortion of the AC line voltage, overcurrents in the neutral line of the three-phase systems and, after all, a poor utilisation of the power system's energy capability. This can be measured in terms of either Total Harmonic Distortion (THD), as norms provides for, or Power Factor (PF), intended as the ratio between the real power (the one transferred to the output) and the apparent power (RMS line voltage times RMS line current) drawn from the mains, which is more immediate. A traditional input stage with capacitive filter has a low PF (0.5-0.7) and a high THD (> 100%). The new European norms and the International standard requirements have spurred the design of high power factor ballasts and they are starting to impose a limit on the input current harmonic content. For these reasons power factor corrector (PFC) is now diffusing in consumer and industrial lighting. With a high power factor switching preregulator, interposed between the input rectifier bridge and the bulk filter capacitor, the power factor will be improved (up to 0.99). The current capability is increased, the bulk capacitor peak current and the harmonic disturbances are reduced. The L6561 is an IC intended to control PFC preregulators by using the transition mode technique and is optimised for lamp ballast applications. The operation can be summarised in the following description (for more information, see AN966). The AC mains voltage, that can range from 85V to 265V, is rectified by a diodes bridge and delivered to the boost converter. The input capacitor has been split in two parts (C1 and C2) to increase the performance in terms of harmonic distortion (THD). In fact, due to the wide range supply and the possibility to change the output power, the minimisation of the capacitor connected after the bridge, allows to reduce the THD. The boost converter consists of a boost inductor (T1), a controlled power switch (Q1), a catch diode (D1), an output capacitor (C7) and, obviously, a control circuitry (see fig.3). The PFC section has been designed to supply a 400V DC and a power of 60W. Ballast Section The regulated voltage is delivered to the ballast section. The ballast is based on the high performances L6574, which is an OFF-LINE half bridge driver designed in 600V BCD technology. It adds to the full integrated half bridge driver topology a built-in voltage controlled oscillator (VCO), a preheating start-up procedure and an operational amplifier dedicated to the feedback loop. To avoid cross conduction of the power MOSFETs or IGBTs, the internal logic ensures a minimum dead time. The load consists of a series resonant circuit (L1-C19) with the lamp connected across the capacitor (C19). This topology allows to operate in Zero Voltage Switching, to reduce the transistor switching losses and the electromagnetic interference generated by the output wiring of the lamp. The blocking capacitor (C17//C18) allows to obtain a zero average lamp current. In steady state the voltage across these capacitors is as high as half the high voltage bus, that is about 200V. Preheating and ignition sequence The turn-on sequence can be divided in three phases: preheating, ignition and normal lamp burning. The preheating of the lamp filaments is achieved by a high switching frequency fPRE, about 60 kHz, set by RPRE = R13 + R14 + R15 and CF = C13, to ensure that a current flows in the filaments without lamp ignition. In fact the initial voltage applied across the lamp is below the strike potential. The duration of the preheating period tPRE is set by the capacitor CPRE = C14. The choice of this time is strictly dependent on the lamp type. In the application tPRE has been set at 1.5 sec. The ignition sequence begins after tPRE. The switching frequency decreases towards the resonance point (L1-
13/20
AN993 APPLICATION NOTE
C19), increasing the voltage across the lamp, and causing the ignition. The time interval in which the frequency shifts, t SH, amounts to tSH = tPRE/10 = 150ms. At the end of tSH the frequency reaches 31 kHz (R19-C13), and then the current feedback loop is activated.
Current feedback loop The current control is achieved by varying the switching frequency of the VCO. Since controlling the average current in the lamp means controlling the output power, it is quite easy to perform dimming function. The OPAMP compares the low-pass filtered half-bridge current, shunted by R25, with a reference, achieved by a partition of the voltage at pin 2 (VPIN2 = 2V). This set-point could be changed by the trimmer R14, to perform the dimming function. The OP-AMP output is connected to RING pin by D3 and R18. The diode D3 is necessary to avoid that the switching frequency decreases below the value set by R19. At start-up the voltage across R25 (fig.14) remains low until the lamp ignition. So the inverting input of OP-AMP (pin 6) stays low too, while the non inverting input (pin 7) is set at a constant voltage (set-point) by the divider R13, R 14 and R15. Therefore the OP-AMP output (pin 5) remains high (5V) until the lamp ignition, and D3 is off. In this condition the L6574 oscillates at fPRE. As the lamp strikes on (after tPRE and tSH), the average voltage across R25 increases and the feedback is able to regulate the lamp current.
Figure 14. Current feedback loop
Q3 STP4NB50/ STP6NB50 C9 8.2nF R33 9.1K R25 0.68
R18 100K D3 1N4148
R19 100K
6
5
7 +
RIGN 4
D98IN818A
Figure 15. Cpre waveform (Ch1) and OP AMP output (Ch2)
14/20
AN993 APPLICATION NOTE
Start-up and supply The start-up procedure is very important in an application that contains two different sections. The ballast section starts before the PFC, avoiding any extra-voltage at the PFC section output, and so the L6561 dynamic OVP activation (see AN966). This behaviour is guaranteed under all conditions because the VS turn-on threshold of L6574 is lower than the L6561 one. At start-up the L6574 is powered by the resistor (R3 + R4). This resistor must be chosen so as to ensure the "before start-up current" of both the L6561 and L6574. When the ballast section is running, the charge pump (C11, R16, D2 and DZ1) allows to supply both the devices. The resistor R16 allows to reduce the peak current. Safety circuitry In normal operation the inductive load ensures a zero voltage switching mode, but if the lamp is disconnected the switching losses in the power MOSFETs will increase considerably. To prevent this occurrence a safety circuitry has been designed. When the lamp is connected the EN1 input (pin 8) of the L6574 is held close to ground by the series of R27, the lamp filament and R25. If the lamp is not present EN1 is pulled up to VS by R26, forcing the L6574 in a latched shutdown state. To resume normal operation it is necessary to turn off the ballast and then turn it on again. A second alarm has been designed to protect the application against the extra voltages which would arise if the lamp did not strike after the ignition sequence, because of an old lamp. A partition of this extra voltage is rectified and delivered to the EN2 input (pin 9) of the L6574, restarting the start-up procedure (preheat and ignition sequence).
Figure 16. Open load safety circuit
+VS
LAMP
C19 8.2nF 1500V
R26 390K EN1 (PIN8)
R25 0.68
D98IN816A
R27 6.8K
C16 1uF
R32 20K
Figure 17. Extra voltage safety circuit
L1 2.1mH C17 100nF 250V C18 100nF 250V LAMP
R28 750K R29 750K D4 1N4148 EN2 (PIN9) R24 6.8K C15 330nF R30 3.9K
C19 8.2nF 1500V
R25 0.68
D98IN817
15/20
AN993 APPLICATION NOTE
DESIGN HINTS Inductance and capacitors evaluation To design an application with L6574 a preliminary evaluation of the components can be done only fixing the lamp type and its electrical characteristics. We can summarise this evaluation process in few marks, but this is an "iterative" process, because we have to do some assumptions that have to be checked at the end of the process. A simplified schematic can well represent the load:
Figure 18. Lamp Simplified Schematic
L L r
lamp
C
rr r
C
Cb
Cb
In fig. 18 r is the lamp filament resistance, rr is the operating lamp equivalent resistance ( when the lamp is off or during preheating rr is an open circuit). First of all we have to evaluate a proper inductance value. L has to give the right current value to the lamp when it is already ignited and is working ("choke" inductance). So it depends on the current required by the lamp , that is to say on the lamp operating wattage and voltage, on the operating frequency fmin, but also on the voltage across L and the lamp (Vb). The greater Vb variation, the greater inductance in order to give a constant current to the lamp. Vb variations are due to the High Voltage Bus variations and to the ripple on half battery capacitor (Cb). The ripple depends on Cb size, and we have to do a proper hypothesis on it to estimate Vb variation. Taking into account all these data and hypothesis, we can do some simplifying assumption: during operating condition the most of the current flows into the lamp, not into C, and all the power delivered to the system is delivered to the lamp. Vlamp is the operating voltage across the lamp and Plamp is the operating lamp wattage, so a good approximation to conduct the choke inductance is:
VL Vla mp ( Vb - Vlamp ) -------- -------------------------------L = -------------------------- ------IL 2 f Pla mp 2 f m in
The second step is the evaluation of the capacitance across the lamp (C). When the lamp is not yet ignited, C has to allow a sufficient current to flow into the lamp filament in order to have a proper preheating. The power to be delivered to the lamp filaments (Pfil ), the preheating frequency, the lamp filament resistance r and the maximum voltage to be applied across C without causing lamp ignition are constraints that help to evaluate the capacitor size. Setting the current through the lamp filament and the max. voltage across the capacitor, you have a capacitor range of values.
1 Pfil 1 -------C = IC -------------------- - ------------- ----------------------------------------------2 f VC r 2 fpre VCm ax
VCmax < Vignition Those values of L, Cb and C have to be corrected in order to have standard and commercial components values. Using these values and lamp equivalent resistances the transfer functions during preheating and operating condition can be calculate.
16/20
AN993 APPLICATION NOTE
The preheating transfer function allows to see if moving towards the resonant frequency (L-C) there is a frequency at which the voltage across the lamp enables its ignition, and this frequency has to be between fmax and fmin. The transfer function gain depends also on r, but r changes greatly during preheating (also 3-4 times) and this has to be considered (see fig.19).
Figure 19. Preheating Transfer Function
Vlamp Vin
F=F(L,C,r@25C)
F=F(L,C,r@Tpre)
freq
1/2sqrt(L C)
fmin
fpre
frequency shifting during Tsh
The operating transfer function allows to check if we have an operating voltage across the lamp (at f = fmin) that is similar to the one used to evaluate L (fig. 20).
Figure 20. Operating Transfer Function
Vlamp Vin
freq
f = f(L,C,rr)
fmin
If one of these condition is not verified, the evaluation process has to be restarted changing the initial hypothesis: i.e. changing the frequencies, or the assumption on Cb ripple. If this everything is OK, the values found can be used , and a preliminary stage of design can be concluded: to better set the components and the frequencies values experiments are needed, also to verify the initial assumptions, and in case to reiterate another step of evaluation process with better assumptions. The following flow chart can help the iteration process:
17/20
AN993 APPLICATION NOTE
Figure 21. Iterative Process
DATA: operating lamp elec.characteristic HVB
HYPOTHESIS: fmin Cb
L evaluation
DATA: ignition and preheating lamp elec.characteristic
HYPOTHESIS: fpre
C evaluation
Transfer preheating funct.
OK
Transfer operating funct.
OK
check on board
OK
18/20
AN993 APPLICATION NOTE
HOW TO DIM THE LAMP In the demo board the lamp is dimmed changing the working frequency. The minimum working frequency is:
1.41 f min = --------------------Rig n C f
To change the working frequency we have to change the current that flows from pin 2. Using fig.9 circuit, the current that flows from pin n. 2 depends also on the op-amp out pin (pin n. 5) voltage: if V5>V2 for sure there will be no current in R18, and the frequency will be equal to fmin. Instead when the op amp out voltage gets lower the current that flows from pin 2 will go into R18 and R19: pin 2 sees the parallel between R19 and an equivalent resistor that depends on V5 and on D3 voltage:
V2 R 18 R18 equ = ----------------------------------------------- V2 - V dio de3 - V5
So the working frequency will be:
1.41 f work = ----------------------------------------------------- ( R ign // R18e qu ) C f
and we have the max fworking when V5=0V For example, in our demo we can calculate a max frequency of ~53KHz (it is V2=2V, and we can assume Vdiode3=0.5V). The dimming level is set changing potentiometer R14 value, that makes change the op amp positive reference from 20-30mV to 110-120mV.If we lower R18 we increase the max working frequency and we can lower the dimming level (higher the frequency, lower the current in the lamp arc). When we try to dim the lamp towards low power range (i.e. <20% electrical arc power) a common effect is the presence of stationary waves along the lamp tube and / or some flickering effects. A common trick to make these disturbs disappear is to add a small continuos current flow inside the lamp (few mA). The easiest way to do it is to add a resistor in parallel to the half battery capacitor (C17 // C18 in fig.9). For instance we can add 50Kohm (I=200V/50Kohm=4mA) and this is already very effective. Of course the resistor has to be able to sustain the power dissipation caused by the current flow, so it is common to use many resistors connected in parallel. Just remember, this is only a trick: it helps but it is not enough. It has to be used together with the right frequencies setting that has to be chosen according to the lamp type.
Dimming level and lamp turn on During the start up sequence the frequency always goes from fmax to fmin, no matter which is the set dimming level. Only after the lamp turn on frequency moves towards higher frequencies. The delay during which f=fmin allows the lamp turn on, but has the drawback of causing a "flash" that can be unpleasant. As the minimum length of this delay depends on the lamp type, it is better to set it as low as possible, finding to best compromise. The easiest way to set this delay is to act on the op amp compensation, i.e. on C9: the higher C9 the longer the delay, the easier the lamp turn on. Values up to 200nF and more are commonly used.
19/20
AN993 APPLICATION NOTE
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. N o license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a registered trademark of STMicroelectronics (R) 2002 STMicroelectronics - All Rights Reserved STMicroelectronics GROUP OF COMPANIES Australia - Brazil - Canada - China - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan -Malaysia - Malta - Morocco Singapore - Spain - Sweden - Switzerland - United Kingdom - United States. http:/ /www.st.com
20/20

▲Up To Search▲

Price & Availability of AN993

	To Download AN993 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .