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rev. d information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. trademarks and registered trademarks are the property of their respective owners. one technology way, p.o. box 9106, norwood, ma 02062-9106, u.s.a. tel: 781/329-4700 www.analog.com fax: 781/326-8703 ? 2004 analog devices, inc. all rights reserved. ad7713 * lc 2 mos loop-powered signal conditioning adc * protected by u.s. patent no. 5,134,401. features charge balancing adc 24 bits no missing codes 0.0015% nonlinearity 3-channel programmable gain front end gains from 1 to 128 2 differential inputs 1 single-ended high voltage input low-pass filter with programmable filter cutoffs ability to read/write calibration coefficients bidirectional microcontroller serial interface single-supply operation low power (3.5 mw typ) with power-down mode (150 w typ) applications loop powered (smart) transmitters rtd transducers process control portable industrial instruments cmos construction ensures low power dissipation, and a hard- ware programmable power-down mode reduces the standby power consumption to only 150 w typical. the part is available in a 24- lead, 0.3 inch wide, pd ip and cerdip as well as a 24- lead soic package. product highlights 1. the ad7713 consumes less than 1 ma in total supply current, making it ideal for use in loop-powered systems. 2. the two programmable gain channels allow the ad7713 to accept input signals directly from a transducer removing a considerable amount of signal conditioning. to maximize the flexibility of the part, the high level analog input accepts 4 v ref signals. on-chip current sources provide excitation for 3-wire and 4-wire rtd configurations. 3. no missing codes ensures true, usable, 24-bit dynamic range coupled with excellent 0.0015% accuracy. the effects of temperature drift are eliminated by on-chip self-calibration, which removes zero-scale and full-scale errors. 4. the ad7713 is ideal for microcontroller or dsp processor applications with an on-chip control register, which allows control over filter cutoff, input gain, signal polarity, and calibration modes. the ad7713 allows the user to read and write the on-chip calibration registers. general description the ad7713 is a complete analog front end for low frequency m easurement applications. the device accepts low level signals directly from a transducer or high level signals (4 v ref ) and outputs a serial digital word. it employs - conversion technique to realize up to 24 bits of no missing codes performance. t he input signal is applied to a proprietary pro- grammable gain front end based around an analog modulator. the modulator output is processed by an on-chip digital filter. the first notch of this digital filter can be programmed via the on-chip control register, allowing adjustment of the filter cutoff and settling time. the part features two differential analog inputs and one single- ended high level analog input as well as a differential reference input. it can be operated from a single supply (av dd and dv dd at 5 v). the part provides two current sources that can be used to provide excitation in 3-wire and 4-wire rtd configurations. the ad7713 thus performs all signal conditioning and conver- sion for a single-, dual- or three-channel system. the ad7713 is ideal for use in smart, microcontroller-based systems. gain settings, signal polarity, and rtd current control can be configured in software using the bidirectional serial port. the ad7713 contains self-calibration, system calibration, and background calibration options and also allows the user to read and to write the on-chip calibration registers. functional block diagram charging balancing adc auto-zeroed - modulator digital filter clock generation ad7713 agnd dgnd mclk out mclk in ain1(+) ain1(?) ain2(+) ref in(+) av dd dv dd av dd 1 a a = 1 ? 128 standby ain2(?) ref in(?) sync rtd2 rtd1 pga av dd 200 a 200 a mux serial interface control register output register mode sdata a0 drdy tfs rfs sclk ain3 input scaling
rev. d e2e ad7713especifications parameter a, s versions 1 unit conditions/comments static performance no missing codes 24 bits min guaranteed by design. for filter notches 12 hz. 22 bits min for filter notch = 20 hz. 18 bits min for filter notch = 50 hz. 15 bits min for filter notch = 100 hz. 12 bits min for filter notch = 200 hz. output noise see tables i and ii depends on filter cutoffs and selected gain. integral nonlinearity 0.0015 % of fsr max filter notches 12 hz; typically 0.0003%. positive full-scale error 2, 3, 4 full-scale drift 5 1 v/ c typ for gains of 1, 2. 0.3 v/ c typ for gains of 4, 8, 16, 32, 64, 128. unipolar offset error 2, 4 unipolar offset drift 5 0.5 v/ c typ for gains of 1, 2. 0.25 v/ c typ for gains of 4, 8, 16, 32, 64, 128. bipolar zero error 2, 4 bipolar zero drift 5 0.5 v/ c typ for gains of 1, 2. 0.25 v/ c typ for gains of 4, 8, 16, 32, 64, 128. gain drift 2 ppm/ c typ bipolar negative full-scale error 2 0.0004 % of fsr max typically 0.0006%. bipolar negative full-scale drift 5 1 v/ c typ for gains of 1, 2. 0.3 v/ c typ for gains of 4, 8, 16, 32, 64, 128. analog inputs input sampling rate, f s see table iii normal-mode 50 hz rejection 6 100 db min for filter notches of 2 hz, 5 hz, 10 hz, 25 hz, 50 hz, 0.02 f notch . normal-mode 60 hz rejection 6 100 db min for filter notches of 2 hz, 6 hz, 10 hz, 30 hz, 60 hz, 0.02 f notch . ain1, ain2 7 input voltage range 8 for normal operation. depends on gain selected. 0 to +v ref 9 v max unipolar input range (b/u bit of control register = 1). v ref v max bipolar input range (b/u bit of control register = 0). common-mode rejection (cmr) 100 db min at dc and av dd = 5 v. 90 db min at dc and av dd = 10 v. common-mode 50 hz rejection 6 150 db min for filter notches of 2 hz, 5 hz, 10 hz, 25 hz, 50 hz, 0.02 f notch. common-mode 60 hz rejection 6 150 db min for filter notches of 2 hz, 6 hz, 10 hz, 30 hz, 60 hz, 0.02 f notch. common-mode voltage range 10 agnd to av dd v min to v max dc input leakage current @ 25 c10 pa max t min to t max 1na max sampling capacitance 6 20 pf max ain3 input voltage range 0 to + 4 v ref v max for normal operation. depends on gain selected. gain error 11 0.05 % typ additional error contributed by resistor attenuator. gain drift 1 ppm/ c typ additional drift contributed by resistor attenuator. offset error 11 4 mv max additional error contributed by resistor attenuator. input impedance 30 k min (av dd = 5 v 5%; dv dd = 5 v 5%; ref in(+) = 2.5 v; ref in(e) = agnd; mclk in = 2 mhz, unless otherwise noted. all specifications t min to t max , unless otherwise noted.)
rev. d ad7713 e3e parameter a, s versions 1 unit conditions/comments reference input ref in(+) e ref in(e) voltage 2.5 to av dd /1.8 v min to v max for specified performance. part is functional with lower v ref voltages. input sampling rate, f s f clk in /512 normal-mode 50 hz rejection 6 100 db min for filter notches of 2 hz, 5 hz, 10 hz, 25 hz, 50 hz, 0.02 f notch . normal-mode 60 hz rejection 6 100 db min for filter notches of 2 hz, 6 hz, 10 hz, 30 hz, 60 hz, 0.02 f notch . common-mode rejection (cmr) 100 db min at dc. common-mode 50 hz rejection 6 150 db min for filter notches of 2 hz, 5 hz, 10 hz, 25 hz, 50 hz, 0.02 f notch . common-mode 60 hz rejection 6 150 db min for filter notches of 2 hz, 6 hz, 10 hz, 30 hz, 60 hz, 0.02 f notch . common-mode voltage range 10 agnd to av dd v min to v max dc input leakage current @ 25 c10 pa max t min to t max 1na max logic inputs input current 10 a max all inputs except mclk in v inl , input low voltage 0.8 v max v inh , input high voltage 2.0 v min mclk in only v inl , input low voltage 0.8 v max v inh , input high voltage 3.5 v min logic outputs v ol , output low voltage 0.4 v max i sink = 1.6 ma. v oh , output high voltage 4.0 v min i source = 100 a. floating state leakage current 10 a max floating state output capacitance 12 9 pf typ transducer burn-out current 1.2 a nom initial tolerance @ 25 c 10 % typ drift 0.1 %/ c typ rtd excitation currents (rtd1, rtd2) output current 200 a nom initial tolerance @ 25 c 20 % max drift 20 ppm/ c typ initial matching @ 25 c 1% ma x matching between rtd1 and rtd2 currents. drift matching 3 ppm/ c typ matching between rtd1 and rtd2 current drift. line regulation (av dd ) 200 na/v max av dd = 5 v. load regulation 200 na/v max system calibration ain1, ain2 positive full-scale calibration limit 13 +(1.05 v ref )/gain v max gain is the selected pga gain (between 1 and 128). negative full-scale calibration limit 13 e(1.05 v ref )/gain v max gain is the selected pga gain (between 1 and 128). offset calibration limit 14, 15 e(1.05 v ref )/gain v max gain is the selected pga gain (between 1 and 128). input span 14 +(0.8 v ref )/gain v min gain is the selected pga gain (between 1 and 128). +(2.1 v ref )/gain v max gain is the selected pga gain (between 1 and 128).
rev. d e4e ad7713 parameter a, s versions 1 unit conditions/comments ain3 positive full-scale calibration limit 13 +(4.2 v ref )/gain v max gain is the selected pga gain (between 1 and 128). offset calibration limit 15 0 to v ref /gain v max gain is the selected pga gain (between 1 and 128). input span +( 3.2 v ref ) /gain v min gain is the selected pga gain (between 1 and 128). +(4.2 v ref )/gain v max gain is the selected pga gain (between 1 and 128). power requirements power supply voltages av dd voltage 16 5 to 10 v nom 5% for specified performance. dv dd voltage 17 5v nom 5% for specified performance. power supply currents av dd current 0.6 ma max av dd = 5 v. 0.7 ma max av dd = 10 v. dv dd current 0.5 ma max f clk in = 1 mhz. digital inputs 0 v to dv dd . 1 ma max f clk in = 2 mhz. digital inputs 0 v to dv dd . power supply rejection 18 rejection w.r.t. agnd. (av dd and dv dd ) 19 db typ power dissipation normal mode 5.5 mw max av dd = dv dd = 5 v, f clk in = 1 mhz; typically 3.5 mw. standby (power-down) mode 300 w max av dd = dv dd = 5 v, typically 150 w. notes 1 temperature range is: a version, e40 c to +85 c; s version, e55 c to +125 c. 2 applies after calibration at the temperature of interest. 3 positive full-scale error applies to both unipolar and bipolar input ranges. 4 these errors will be of the order of the output noise of the part as shown in table i after system calibration. these errors wi ll be 20 v typical after self-calibration or background calibration. 5 recalibration at any temperature or use of the background calibration mode will remove these drift errors. 6 these numbers are guaranteed by design and/or characterization. 7 the ain1 and ain2 analog inputs present a very high impedance dynamic load that varies with clock frequency and input sample ra te. the maximum recom- mended source resistance depends on the selected gain. 8 the analog input voltage range on the ain1(+) and ain2(+) inputs is given here with respect to the voltage on the ain1(e) and a in2(e) inputs. the input voltage range on the ain3 input is with respect to agnd. the absolute voltage on the ain1 and ain2 inputs should not go more positive t han av dd + 30 mv or more negative than agnd e 30 mv. 9 v ref = ref in(+) e ref in(e). 10 this common-mode voltage range is allowed, provided that the input voltage on ain(+) and ain(e) does not exceed av dd + 30 mv and agnd e 30 mv. 11 this error can be removed using the system calibration capabilities of the ad7713. this error is not removed by the ad7713?s se lf-calibration feature. the offset drift on the ain3 input is four times the value given in the static performance section of the specifications. 12 guaranteed by design, not production tested. 13 after calibration, if the analog input exceeds positive full scale, the converter will output all 1s. if the analog input is le ss than negative full scale, the device will output all 0s. 14 these calibration and span limits apply provided the absolute voltage on the ain1 and ain2 analog inputs does not exceed av dd + 30 mv or go more negative than agnd e 30 mv. 15 the offset calibration limit applies to both the unipolar zero point and the bipolar zero point. 16 operating with av dd voltages in the range 5.25 v to 10.5 v is guaranteed only over the 0 c to 70 c temperature range. 17 the 5% tolerance on the dv dd input is allowed provided that dv dd does not exceed av dd by more than 0.3 v. 18 measured at dc and applies in the selected pass band. psrr at 50 hz will exceed 120 db with filter notches of 2 hz, 5 hz, 10 hz , 25 hz, or 50 hz. psrr at 60 hz will exceed 120 db with filter notches of 2 hz, 6 hz, 10 hz, 30 hz, or 60 hz. 19 psrr depends on gain: gain of 1 = 70 db typ; gain of 2 = 75 db typ; gain of 4 = 80 db typ; gains of 8 to 128 = 85 db typ. specifications subject to change without notice.
rev. d ad7713 e5e timing characteristics 1, 2 (dv dd = 5 v 5%; av dd = 5 v or 10 v 5%; agnd = dgnd = 0 v; f clkin = 2 mhz; input logic 0 = 0 v, logic 1 = dv dd , unless otherwise noted.) limit at t min , t max parameter (a, s versions) unit conditions/comments f clk in 3, 4 400 khz min master clock frequency: crystal oscillator or 2 mhz max externally supplied for specified performance t clk in lo 0.4 t clk in ns min master clock input low time; t clk in = 1/f clk in t clk in hi 0.4 t clk in ns min master clock input high time t r 5 50 ns max digital output rise time; typically 20 ns t f 5 50 ns max digital output fall time; typically 20 ns t 1 1000 ns min sync sc 2 drdy rfs st drdy rfs t 2 cn a rfs st a rfs t cn 2 rfs scf cn 2 dat rfs d cn 2 scfdd cn 2 cn 2 sc cn 2 sc a tfs st a tfs t cn 2 tfs scfdt cn tfs scft dscst dsct c sc cn scf 2 drdy rfs st 2 drdy rfs t 22 2 cn a rfs st 2 a rfs t 2 cn dat rfs d 2 scfdd 2 cn 2 2 2 cn sc 2 2 cn sc 2 cn scf drdy 2 scdt cn rfs tfs scft cn 2 rfs dt 2 a tfs st a tfs t cn scf tfs t 2 cn sc dscst dsct nts a 2 sf cncnad tad cn 2 s tf 2 tf t ft
rev. d e6e ad7713 caution esd (electrostatic discharge) sensitive device. electrostatic charges as high as 4000 v readily accumulate on the human body and test equipment and can discharge without detection. although the ad7713 features proprietary esd protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. therefore, proper esd precautions are recommended to avoid performance degradation or loss of functionality. 2.1v 1.6ma 200 a to output pin 100pf figure 1. load circuit for access time and bus relinquish time absolute maximum ratings * (t a = 25 c, unless otherwise noted.) av dd to agnd . . . . . . . . . . . . . . . . . . . . . . . e0.3 v to +12 v av dd to dgnd . . . . . . . . . . . . . . . . . . . . . . . e0.3 v to +12 v dv dd to agnd . . . . . . . . . . . . . . . . . . . . . . . . e0.3 v to +6 v dv dd to dgnd . . . . . . . . . . . . . . . . . . . . . . . . e0.3 v to +6 v agnd to dgnd . . . . . . . . . . . . . . . . . . . . . . . e0.3 v to +6 v ain1, ain2 input voltage to agnd . . . . . . . . . . . . . . . . . . . . . e0.3 v to av dd + 0.3 v ain3 input voltage to agnd . . . . . . . . . . . . e0.3 v to +22 v reference input voltage to agnd . . . e0.3 v to av dd + 0.3 v digital input voltage to dgnd . . . . . e0.3 v to av dd + 0.3 v digital output voltage to dgnd . . . e0.3 v to dv dd + 0.3 v operating temperature range commercial (a version) . . . . . . . . . . . . . . . e40 c to +85 c extended (s version) . . . . . . . . . . . . . . . . . e55 c to +125 c storage temperature range . . . . . . . . . . . . . e65 c to +150 c junction temperature . . . . . . . . . . . . . . . . . . . . . . . . . . 150 c pdip package, power dissipation . . . . . . . . . . . . . . . . 450 mw ja thermal impedance . . . . . . . . . . . . . . . . . . . . . . 105 c/w lead temperature, soldering (10 sec) . . . . . . . . . . . . 260 c cerdip package, power dissipation . . . . . . . . . . . . . 450 mw ja thermal impedance . . . . . . . . . . . . . . . . . . . . . . . 70 c/w lead temperature, soldering . . . . . . . . . . . . . . . . . . 300 c soic package, power dissipation . . . . . . . . . . . . . . . . 450 mw ja thermal impedance . . . . . . . . . . . . . . . . . . . . . . . 75 c/w lead temperature, soldering vapor phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . . 215 c infrared (15 secs) . . . . . . . . . . . . . . . . . . . . . . . . . . 220 c power dissipation (any package) to 75 c . . . . . . . . . . 450 mw * stresses above those listed under absolute maximum ratings may cause perma- nent damage to the device. this is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of the specification is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. ordering guide temperature model range package option * ad7713an e40 c to +85 c n-24 ad7713ar e40 c to +85 c rw-24 ad7713ar-reel e40 c to +85 c rw-24 ad7713ar-reel7 e40 c to +85 c rw-24 ad7713aq e40 c to +85 c q-24 ad7713sq e55 c to +125 c q-24 * n = pdip; q = cerdip; rw = soic.
rev. d ad7713 e7e pin configuration pdip, cerdip, and soic 1 sclk 2 mclk in 3 mclk out 4 a0 5 sync 6 mode 7 ain1(+) 10 ain2(?) 9 ain2(+) 8 ain1(?) 11 standby 12 av dd 24 dgnd 23 dv dd 22 sdata 21 drdy 20 rfs 19 tfs 18 agnd 15 ref in(+) 16 rtd2 17 ain3 14 ref in(?) 13 rtd1 ad7713 top view (not to scale) pin function description pin no. mnemonic function 1 sclk serial clock. logic input/output, depending on the status of the mode pin. when mode is high, the device is in its self-clocking mode, and the sclk pin provides a serial clock output. this sclk be- comes active when rfs tfs rfs tfs d sct a ad 2 cn csdta cnctacn csctt 2 ct cnct a a sync aad d an a ct an t an acn an2 ac2 an2 ac2n standby t 2 a dd as rtd cca2 a rtdt rfn rtrfna dd andrfn rfn rfn rtrfnrf nrfna dd and
rev. d e8e ad7713 pin no. mnemonic function 16 rtd2 constant current output. a nominal 200 a constant current is provided at this pin, which can be used as the excitation current for rtds. this current can be turned on or off via the control register. this second current can be used to eliminate lead resistanced errors in 3-wire rtd configurations. 17 ain3 analog input channel 3. high level analog input that accepts an analog input voltage range of 4 v ref /gain. at the nominal v ref of 2.5 v and a gain of 1, the ain3 input voltage range is 0 v to 10 v. 18 agnd ground reference point for analog circuitry. 19 tfs tfsa tfs tfs 2 rfs rfsa scsdata rfs sdata rfs 2 drdy at drdy drdy ad 22 sdata sd d rfs drdy d sc tfs t 2 d dd dsd dd a dd 2 dnd rdc trny n t t sb sb t fs fanan2 an rf an2sban anan2an rf an2sb fanan2 ansban sb b t fan an2ansban bnfs t fanan2 an rf ansban fs anan2 an rf anan rf an nfs t anan2an rf an cr ad t ad fscr tad s adt ad
rev. d ad7713 e9e control register (24 bits) a write to the device with the a0 input low writes data to the control register. a read to the device with the a0 input low accesses the contents of the control register. the control register is 24 bits wide. when writing to the register, 24 bits of data must be written; otherwise, the data will not be loaded to the msb md2 md1 md0 g2 g1 g0 ch1 ch0 wl ro bo b/u fs11 fs10 fs9 fs8 fs7 fs6 fs5 fs4 fs3 fs2 fs1 fs0 lsb operating mode md2 md1 md0 operating mode 00 0 normal mode. this is the normal mode of operation of the device whereby a read to the device with a0 high accesses data from the data register. this is the default condition of these bits after the internal power-on reset. 00 1a ctivate self-calibration. this activates self-calibration on the channel selected by ch0 and ch1. this is a 1-step calibration sequence, and when complete, the part returns to normal mode (with md2, md1, md0 of the control registers returning to 0, 0, 0). the drdy f rf a sctcc t2 drdy t 2 a sct drdy asctc ct drdy f rf abctc cad t rf r sccaa cca acc t2 t2 r fsccaa cca acct 2 t2 2 2 tfs 2s 2
rev. d e10e ad7713 pga gain g2 gl g0 gain 00 01 (default condition after the internal power-on reset) 0012 0104 0118 10016 10132 11064 111128 channel selection ch1 ch0 channel 00 ain1 (default condition after the internal power-on reset) 01 ain2 10 ain3 word length wl output word length 0 16-bit (default condition after the internal power-on reset) 1 24-bit rtd excitation currents ro 0o ff (default condition after the internal power-on reset) 1on burn-out current bo 0o ff (default condition after the internal power-on reset) 1on bipolar/unipolar selection (both inputs) b/u 0 bipolar (default condition after the internal power-on reset) 1 unipolar filter selection (fs11 to fs0) the on-chip digital filter provides a sinc 3 (or (sinx/x) 3 ) filter response. the 12 bits of data programmed into these bits deter- mine the filter cutoff frequency, the position of the first notch of the filter, and the data rate for the part. in association with the gain selection, it also determines the output noise (and therefore the effective resolution) of the device. the first notch of the filter occurs at a frequency determined by the relationship: filter first notch frequency = ( f clk in /512)/ code where code is the decimal equivalent of the code in bits fs0 to fs11 and is in the range 19 to 2,000. with the nominal f clk in of 2 mhz, this results in a first notch frequency range from 1.952 hz to 205.59 khz. to ensure correct operation of the ad7713, the value of the code loaded to these bits must be within this range. failure to do this will result in unspecified operation of the device. changing the filter notch frequency, as well as the selected gain, impacts resolution. tables i and ii and figures 2a and 2b show the effect of the filter notch frequency and gain on the effective resolution of the ad7713. the output data rate (or effective conversion time) for the device is equal to the frequency se- lected for the first notch of the filter. for example, if the first notch of the filter is selected at 10 hz, then a new word is avail- able at a 10 hz rate or every 100 ms. if the first notch is at 200 hz, a new word is available every 5 ms. the settling time of the filter to a full-scale step input change is worst case 4 1/( o utput data rate ). this settling time is to 100% of the final value. for example, with the first filter notch at 100 hz, the settling time of the filter to a full-scale step input change is 400 ms max. if the first notch is at 200 hz, the settling time of the filter to a full-scale input step is 20 ms max. this settling time can be reduced to 3 l/( o utput data rate ) by synchronizing the step input change to a reset of the digital filter. in other words, if the step input takes place with sync output data rate ). if a change of channels takes place, the settling time is 3 l/( output data rate ) regardless of the sync tb filter db frequency first notch frequency ?= 30 262 .
rev. d ad7713 e11e table i. output noise vs. gain and first notch frequency first notch of typical output rms noise ( v) filter and o/p e3 db data rate 1 f re que nc yg ain of 1 gain of 2 gain of 4 gain of 8 g ain of 16 gain of 32 gain of 64 gain of 128 2 hz 2 0.52 hz 1.0 0.78 0.48 0.33 0.25 0.25 0.25 0.25 5 hz 2 1.31 hz 1.8 1.1 0.63 0.5 0.44 0.41 0.38 0.38 6 hz 2 1.57 hz 2.5 1.31 0.84 0.57 0.46 0.43 0.4 0.4 10 hz 2 2.62 hz 4.33 2.06 1.2 0.64 0.54 0.46 0.46 0.46 12 hz 2 3.14 hz 5.28 2.36 1.33 0.87 0.63 0.62 0.6 0.56 20 hz 3 5.24 hz 13 6.4 3.7 1.8 1.1 0.9 0.65 0.65 50 hz 3 13.1 hz 130 75 25 12 7.5 4 2.7 1.7 100 hz 3 26.2 hz 0.6 10 3 0.26 10 3 140 70 35 25 15 8 200 hz 3 52.4 hz 3.1 10 3 1.6 10 3 0.7 10 3 0.29 10 3 180 120 70 40 notes 1 the default condition (after the internal power-on reset) for the first notch of filter is 60 hz. 2 for these filter notch frequencies, the output rms noise is primarily dominated by device noise, and, as a result, is independe nt of the value of the reference voltage. therefore, increasing the reference voltage will give an increase in the effective resolution of the device (i.e., the ratio of the rms noise to the input full scale is increased since the output rms noise remains constant as the input full scale increases). 3 for these filter notch frequencies, the output rms noise is dominated by quantization noise, and, as a result, is proportional to the value of the reference voltage. table ii. effective resolution vs. gain and first notch frequency first notch of effective resolution * (bits) filter and o/p e3 db data rate frequency gain of 1 gain of 2 gain of 4g ain of 8 gain of 16 gain of 32 gain of 64 gain of 128 2 hz 0.52 hz 22.5 21.5 21.5 21 20.5 19.5 18.5 17.5 5 hz 1.31 hz 21.5 21 21 20 19.5 18.5 17.5 16.5 6 hz 1.57 hz 21 21 20.5 20 19.5 18.5 17.5 16.5 10 hz 2.62 hz 20 20 20 19.5 19 18.5 17.5 16.5 12 hz 3.14 hz 20 20 20 19.5 19 18 17 16 20 hz 5.24 hz 18.5 18.5 18.5 18.5 18 17.5 17 16 50 hz 13.1 hz 15 15 15.5 15.5 15.5 15.5 15 14.5 100 hz 26.2 hz 13 13 13 13 13 12.5 12.5 12.5 200 hz 52.4 hz 10.5 10.5 11 11 11 10.5 10 10 * effective resolution is defined as the magnitude of the output rms noise with respect to the input full scale (i.e., 2 v ref /gain). table ii applies for a v ref of 2.5 v and resolution numbers are rounded to the nearest 0.5 lsb. tables i and ii show the output rms noise for some typical notch and e3 db frequencies. the numbers given are for the bipolar input ranges with a v ref of 2.5 v. these numbers are typical and are generated with an analog input voltage of 0 v. the output noise from the part comes from two sources. first, there is the electrical noise in the semiconductor devices used in the implementation of the m odulator (device noise). second, when the analog input signal is converted into the digital domain, quantization noise is added. t he device noise is at a low level and is largely independent of frequency. the quantization noise starts at an even lower level but rises rapidly with increasing frequency to become the dominant noise source. consequently, lower filter notch settings (below 12 hz approximately) tend to be device noise dominated while higher notch settings are domi- nated by quantization noise. changing the filter notch and cutoff frequency in the quantization noise dominated region results in a more dramatic improvement in noise performance than it does in the device noise dominated region as shown in t able i. furth ermore, quantization noise is added after the pga, so effective resolution is independent of gain for the higher filter notch frequencies. meanwhile, device noise is added in the pga and, therefore, effective resolution suffers a little at high gains for lower notch frequencies. at the lower filter notch settings (below 12 hz), the no missing codes performance of the device is at the 24-bit level. at the higher settings, more codes will be missed until at 200 hz notch setting, no missing codes performance is guaranteed only to the 12-bit level. however, since the effective resolution of the part is 10.5 bits for this filter notch setting; this no missing codes performance should be more than adequate for all applications. the effective resolution of the device is defined as the ratio of the output rms noise to the input full scale. this does not remain constant with increasing gain or with increasing bandwidth. table ii is the same as table i except that the output is expressed in terms of effective resolution (the magnitude of the rms noise with respect to 2 v ref /gain, i.e., the input full scale). it is possible to do post filtering on the device to improve the output data rate for a given e3 db frequency and also to further reduce the output noise (see the digital filtering section).
rev. d ?2 ad7713 figures 2a and 2b gives similar information to that outlined in table i. in this plot, the output rms noise is shown for the full range of available cutoff frequencies rather than for some typical cutoff frequencies as in tables i and ii. the numbers given in these plots are typical values at 25 c. gain of 1 gain of 2 gain of 4 gain of 8 0.1 1.0 1000.0 100.0 10.0 10000.0 10 100 1k 10 k notch frequency (hz) output noise ( v) figure 2a. plot of output noise vs. gain and notch frequency (gains of 1 to 8) gain of 16 gain of 32 gain of 64 gain of 128 0.1 1.0 100.0 10.0 1000.0 10 100 1k 10 k notch frequency (hz) output noise ( v) figure 2b. plot of output noise vs. gain and notch frequency (gains of 16 to 128) circuit description the ad7713 is a - adc with on-chip digital filtering, intended for the measurement of wide dynamic range, low frequency signals, such as those in industrial control or process control applications. it contains a - (or charge balancing) adc, a calibration microcontroller with on-chip static ram, a clock oscillator, a digital filter, and a bidirectional serial communications port. the part contains three analog input channels, two program- mable gain differential input channels, and one programmable gain high-level single-ended input channel. the gain range on both inputs is from 1 to 128. for the ain1 and ain2 inputs, this means that the input can accept unipolar signals of between 0 mv to 20 mv and 0 v to 2.5 v or bipolar signals in the range from 20 mv to 2.5 v when the reference input voltage equals 2.5 v. the input voltage range for the ain3 input is 4 v ref / gain and is 0 v to 10 v with the nominal reference of 2.5 v and a analog gain of 1. the input signal to the selected analog input channel is continuously sampled at a rate determined by the frequency of the master clock, mclk in, and the selected gain (see tabl e iii). a charge balancing adc ( - modulator) converts the sampled signal into a digital pulse train whose duty cycle contains the digital information. the programmable gain function on the analog input is also incorporated in this - modulator with the input sampling frequency being modified to give the higher gains. a sinc 3 digital low-pass filter processes the output of the - modulator and updates the output register at a rate determined by the first notch frequency of this filter. the output data can be read from the serial port randomly or peri- odically at any rate up to the output register update rate. the first notch of this digital filter (and therefore its ? db frequency) can be programmed via an on-chip control register. the programmable range for this first notch frequency is from 1.952 hz to 205.59 hz, giving a programmable range for the ? db frequency of 0.52 hz to 53.9 hz. the basic connection diagram for the part is shown in figure 3. this shows the ad7713 in the external clocking mode with both the av dd and dv dd pins of the ad7713 being driven from the analog 5 v supply. some applications will have sepa- rate supplies for both av dd and dv dd , and in some of these cases, the analog supply will exceed the 5 v digital supply (see the power supplies and grounding section). ref in(+) ain1(+) ain1(?) ain3 av dd dv dd agnd dgnd mclk in mclk out ref in(?) analog 5v supply 0.1 f 10 f ad7713 differential analog input single-ended analog input analog ground digital ground mode dv dd standby dv dd ain2(+) ain2(?) 2.5v reference drdy data ready tfs transmit (write) rfs receive (read) sdata serial data sclk serial clock a0 address input differential analog input sync figure 3. basic connection diagram the ad7713 provides a number of calibration options that can be programmed via the on-chip control register. a calibration cycle can be initiated at any time by writing to this control regis- ter. the part can perform self-calibration using the on-chip calibration microcontroller and sram to store calibration pa ra met e rs. other system components may also be included in the calibration loop to remove offset and gain errors in the input channel using the system calibration mode. another option is a background calibration mode where the part continuously performs self-calibration and updates the calibration coeffi- ci en ts. once the part is in this mode, the user does not have to worry about issuing periodic calibration commands to the device or asking the device to recalibrate when there is a change in the ambient temperature or power supply voltage.
rev. d ad7713 ?3 the ad7713 gives the user access to the on-chip calibration registers, allowing the microprocessor to read the device? calibra- tion coefficients and also to write its own calibration coefficients to the part from prestored values in e 2 prom. this gives the microprocessor much greater control over the ad7713? calibra- tion procedure. it also means that the user can verify that the device has performed its calibration correctly by comparing the coefficients after calibration with prestored values in e 2 prom. for battery-operated or low power systems, the ad7713 offers a standby mode (controlled by the standby pin) that reduces idle power consumption to typically 150 w. theory of operation the general block diagram of a - adc is shown in figure 4. it contains the following elements: ? sample-hold amplifier ? differential amplifier or subtracter ? n analog low-pass filter ? 1-bit adc (comparator) ? 1-bit dac analog low-pass filter dac digital filter digital data comparator s/h amp figure 4. general - adc in operation, the analog signal sample is fed to the subtracter, along with the output of the 1-bit dac. the filtered difference signal is fed to the comparator, whose output samples the differ- ence signal at a frequency many times that of the analog signal sampling frequency (oversampling). oversampling is fundamental to the operation of - adcs. using the quantization noise formula for an adc snr number of bits db = + () 602 176 .. a 1-bit adc or comparator yields an snr of 7.78 db. the ad7713 samples the input signal at a frequency of 7.8 khz or greater (see table iii). as a result, the quantization noise is spread over a much wider frequency than that of the band of interest. the noise in the band of interest is reduced still fur- ther by analog filtering in the modulator loop, which shapes the quantization noise spectrum to move most of the noise energy to frequencies outside the bandwidth of interest. the noise performance is thus improved from this 1-bit level to the performance outlined in tables i and ii and in figures 2a and 2b. the output of the comparator provides the digital input for the 1-bit dac, so that the system functions as a negative feedback loop that tries to minimize the difference signal. the digital data that represents the analog input voltage is contained in the duty cycle of the pulse train appearing at the output of the compara- tor. it can be retrieved as a parallel binary data-word using a digital filter. - adcs are generally described by the order of the analog low-pass filter. a simple example of a first-order - adc is shown in figure 5. this contains only a first-order low-pass filter or integrator. it also illustrates the derivation of the alter- native name for these devices: charge balancing adcs. comparator differential amplifier integrator +fs ?fs dac v in figure 5. basic charge-balancing adc it consists of a differential amplifier (whose output is the differ- ence between the analog input and the output of a 1-bit dac), an integrator, and a comparator. the term charge balancing comes from the fact that this system is a negative feedback loop that tries to keep the net charge on the integrator capacitor at 0 by balanc- ing charge injected by the input voltage with charge injected by the 1-bit dac. when the analog input is 0, the only contribu- tion to the integrator output comes from the 1-bit dac. for the net charge on the integrator capacitor to be 0, the dac output must spend half its time at +fs and half its time at ?s. assum- ing ideal components, the duty cycle of the comparator will be 50%. when a positive analog input is applied, the output of the 1-bit dac must spend a larger proportion of the time at +fs, so the duty cycle of the comparator increases. when a negative input voltage is applied, the duty cycle decreases. the ad7713 uses a second-order - modulator and a digital filter that provides a rolling average of the sampled output. after power-up or if there is a step change in the input voltage, there is a settling time that must elapse before valid data is obtained. input sample rate the modulator sample frequency for the device remains at f clk in /512 (3.9 khz @ f clk in = 2 mhz) regardless of the selected gain. ho w ever, gains greater than 1 are achieved by a combination of multiple input samples per modulator cycle and a scaling of the ratio of the reference capacitor to input capaci- tor. as a result of the multiple sampling, the input the sample rate of the device varies with the selected gain (see table iii). the effective input impedance is 1/ c f s , where c is the input sampling capacitance and f s is the input sample rate. table iii. input sampling frequency vs. gain gain input sampling frequency (f s ) 1f clk in /256 (7.8 khz @ f clk in = 2 mhz) 22 f clk in /256 (15.6 khz @ f clk in = 2 mhz) 44 f clk in /256 (31.2 khz @ f clk in = 2 mhz) 88 f clk in /256 (62.4 khz @ f clk in = 2 mhz) 16 8 f clk in /256 (62.4 khz @ f clk in = 2 mhz) 32 8 f clk in /256 (62.4 khz @ f clk in = 2 mhz) 64 8 f clk in /256 (62.4 khz @ f clk in = 2 mhz) 128 8 f clk in /256 (62.4 khz @ f clk in = 2 mhz)
rev. d ?4 ad7713 digital filtering the ad7713? digital filter behaves like a similar analog filter, with a few minor differences. first, since digital filtering occurs after the a-to-d conversion process, it can remove noise injected during the conversion process. analog filtering cannot do this. on the other hand, analog filtering can remove noise superimposed on the analog signal before it reaches the adc. digital filtering cannot do this, and noise peaks riding on signals near full scale have the potential to saturate the analog modulator and digital filter, even though the average value of the signal is within limits. to alleviate this problem, the ad7713 has overrange headroom built into the - modulator and digital filter, which allows over- range excursions of 5% above the analog input range. if noise signals are larger than this, consideration should be given to analog input filtering or to reducing the input channel voltage so that its full scale is half that of the analog input channel full scale. this will provide an overrange capability greater than 100% at the expense of reducing the dynamic range by 1 bit (50%). filter characteristics the cutoff frequency of the digital filter is determined by the value loaded to bits fs0 to fs11 in the control register. at the maxi- mum clock frequency of 2 mhz, the minimum cutoff frequency of the filter is 0.52 hz, while the maximum programmable cutoff frequency is 53.9 hz. figure 6 shows the filter frequency response for a cutoff frequency of 0.52 hz, which corresponds to a first filter notch frequency of 2 h z. this is a (sinx/x) 3 response (also called sinc 3 ) that provides >100 db of 50 hz and 60 hz rejection. programming a differ- ent cutoff frequency via fs0 to fs11 does not alter the profile of the filter response; it changes the frequency of the notches as outlined in the control register section. ?240 ?180 ?200 ?220 ?60 ?80 ?100 ?120 ?140 ?160 0 ?20 ?40 024681012 frequency (hz) gain (db) figure 6. frequency response of ad7713 filter since the ad7713 contains this on-chip, low-pass filtering, there is a settling time associated with step function inputs, and data on the output will be invalid after a step change until the settling time has elapsed. the settling time depends upon the notch frequency chosen for the filter. the output data rate equates to this filter notch frequency, and the settling time of the filter to a full-scale step input is four times the output data period. in applications using both input channels, the settling time of the filter must be allowed to elapse before data from the second channel is accessed. post filtering the on-chip modulator provides samples at a 3.9 khz output rate. the on-chip digital filter decimates these samples to provide data at an output rate that corresponds to the pro- grammed first notch frequency of the filter. since the output data rate exceeds the nyquist criterion, the output rate for a given bandwidth will satisfy most application requirements. however, there may be some applications that require a higher data rate for a given bandwidth and noise performance. appli- cations that need this higher data rate will require some post filtering following the digital filter of the ad7713. for example, if the required bandwidth is 1.57 hz but the required update rate is 20 hz, the data can be taken from the ad7713 at the 20 hz rate giving a 3 db bandwidth of 5.24 hz. post filtering can be applied to this to reduce the bandwidth and output noise, to the 1.57 hz bandwidth level, while maintaining an output rate of 20 hz. post filtering can also be used to reduce the output noise from the device for bandwidths below 0.52 hz. at a gain of 128, the output rms noise is 250 nv. this is essentially device noise or white noise, and since the input is chopped, the noise has a flat frequency response. by reducing the bandwidth below 0.52 hz, the noise in the resultant pass band can be reduced. a reduction in bandwidth by a factor of 2 results in a 2 reduction in the output rms noise. this additional filtering will result in a longer settling time. antialias considerations the digital filter does not provide any rejection at integer multiples of the modulator sample frequency (n 3.9 khz, where n = 1, 2, 3...). this means that there are frequency bands, f 3 db wide (f 3 db is cutoff frequency selected by fs0 to fs11), where noise passes unattenuated to the output. however, due to the ad7713? high oversampling ratio, these bands occupy only a small fraction of the spectrum, and most broadband noise is filtered. in any case, because of the high oversampling ratio, a simple, rc, single-pole filter is generally sufficient to attenuate the signals in these bands on the analog input and thus provide adequate antialiasing filtering. if passive components are placed in front of the ain1 and ain2 inputs of the ad7713, care must be taken to ensure that the source impedance is low enough so as not to introduce gain errors in the system. the dc input impedance for the ain1 and ain2 inputs is over 1 g ? . the input appears as a dynamic load that varies with the clock frequency and with the selected gain (see figure 7). the input sample rate, as shown in table iii, determines the time allowed for the analog input capacitor, c in , to be charged. external impedances result in a longer charge time for this capacitor, which result in gain er- rors being introduced on the analog inputs. both inputs of the differential input channels look into similar input circuitry. ain high impedance > 1g r int (7k typ) c int (11.5pf typ) switching frequency depends on f clkin and selected gain v bias figure 7. ain1, ain2 input impedance
rev. d ad7713 ?5 in any case, the error introduced due to longer charging times is a gain error that can be removed using the system calibration capabilities of the ad7713 provided that the resultant span is within the span limits of the system calibration techniques for the ad7713. the ain3 input contains a resistive attenuation network as outlined in figure 8. the typical input impedance on this input is 44 k ? . as a result, the ain3 input should be driven from a low impedance source. ain3 33k 11k v bias modulator circuit figure 8. ain3 input impedance analog input functions analog input ranges the analog inputs on the ad7713 provide the user with consid- erable flexibility in terms of analog input voltage ranges. two of the inputs are differential, programmable-gain, input channels that can handle either unipolar or bipolar input signals. the common-mode range of these inputs is from agnd to av dd , provided that the absolute value of the analog input voltage lies between agnd ?30 mv and av dd + 30 mv. the third analog input is a single-ended, programmable gain high-level input that accepts analog input ranges of 0 to 4 v ref /gain. the dc input leakage current on the ain1 and ain2 inputs is 10 pa maximum at 25 c ( 1 na over temperature). this results in a dc offset voltage developed across the source impedance. how ever, this dc offset effect can be compensated for by a combination of the differential input capability of the part and its system calibration mode. the dc input current on the ain3 input depends on the input voltage. for the nominal input voltage range of 10 v, the input current is 225 a typ. burn out current the ain1(+) input of the ad7713 contains a 1 a current source that can be turned on/off via the control register. this current source can be used in checking that a transducer has not burnt out or gone open circuit before attempting to take measurements on that channel. if the current is turned on and is allowed flow into the transducer and a measurement of the input voltage on the ain1 input is taken, it can indicate that the transducer is not func- tioning correctly. for normal operation, this burn out current is turned off by writing a 0 to the bo bit in the control register. rtd excitation currents the ad7713 also contains two matched 200 a constant cur- rent sources which are provided at the rtd1 and rtd2 pins of the device. these currents can be turned on/off via the control register. writing a 1 to the ro bit of the control register enables these excitation currents. for 4-wire rtd applications, one of these excitation currents is used to provide the excitation current for the rtd; the second current source can be left unconnected. for 3-wire rtd con- figurations, the second on-chip current source can be used to eliminate errors due to voltage drops across lead resistances. figures 19 and 20 in the application section show some rtd configurations with the ad7713. the temperature coefficient of the rtd current sources is typically 20 ppm/ c with a typical matching between the temperature coefficients of both current sources of 3 ppm/ c. for applications where the absolute value of the temperature coefficient is too large, the following schemes can be used to remove the drift error. the conversion result from the ad7713 is ratiometric to the v ref voltage. therefore, if the v ref voltage varies with the rtd temperature coefficient, the temperature drift from the current source will be removed. for 4-wire rtd applications, the refer- ence voltage can be made ratiometric to the rtd current source by using the second current with a low tc resistor to generate the reference voltage for the part. in this case, if a 12.5 k ? resistor is used, the 200 a current source generates 2.5 v across the resistor. this 2.5 v can be applied to the ref in(+) input of the ad7713 and the ref in(? input at ground will supply a v ref of 2.5 v for the part. for 3-wire rtd configurations, the reference voltage for the part is generated by placing a low tc resistor (12.5 k ? for 2.5 v reference) in series with one of the constant current sources. the rtd current sources can be driven to within 2 v of av dd . the reference input of the ad7713 is differential so the ref in(+) and ref in(? of the ad7713 are driven from either side of the resistor. both schemes ensure that the reference voltage for the part tracks the rtd current sources over temperature and, thereby, removes the temperature drift error. bipolar/unipolar inputs two analog inputs on the ad7713 can accept either unipolar or bipolar input voltage ranges while the third channel accepts only unipolar signals. bipolar or unipolar options for ain1 and ain2 are chosen by programming the b/u bit of the control register. this programs both channels for either unipolar or bipolar opera- tion. programming the part for either unipolar or bipolar operation does not change any of the input signal conditioning; it simply changes the data output coding. the data coding is binary for unipolar inputs and offset binary for bipolar inputs. the ain1 and ain2 input channels are differential, and as a result, the voltage to which the unipolar and bipolar signals are referenced is the voltage on the ain1() and ain2() inputs. for example, if ain1() is 1.25 v and the ad7713 is configured for unipolar operation with a gain of 1 and a v ref of 2.5 v, the input voltage range on the ain1(+) input is 1.25 v to 3.75 v. for the ain3 input, the input signals are referenced to agnd. reference input the reference inputs of the ad7713, ref in(+) and ref in(?, provide a differential reference input capability. the common- mode range for these differential inputs is from v ss to av dd . the nominal differential voltage, v ref (ref in(+) ?ref in(?), is 2.5 v for specified operation, but the reference voltage can go to 5 v with no degradation in performance, provided that the absolute value of ref in(+) and ref in(? does not exceed its av dd and agnd limits. the part is also functional with v ref voltages down to 1 v, but with degraded performance as the output noise will, in terms of lsb size, be larger. ref in(+) must always be greater than ref in(? for correct operation of the ad7713. both reference inputs provide a high impedance, dynamic load similar to the analog inputs. the maximum dc input leakage cur- rent is 10 pa ( 1 na over temperature), and source resistance may result in gain errors on the part. the reference inputs
rev. d ?6 ad7713 look like the ain1 analog input (see figure 7). in this case, r int is 5 k ? typ and c int varies with gain. the input sample rate is f clk in /256 and does not vary with gain. for gains of 1 to 8, c int is 20 pf; for a gain of 16, it is 10 pf; for a gain of 32, it is 5 pf; for a gain of 64, it is 2.5 pf; and for a gain of 128, it is 1.25 pf. the digital filter of the ad7713 removes noise from the reference input just as it does with the analog input, and the same limita- tions apply regarding lack of noise rejection at integer multiples of the sampling frequency. the output noise performance outlined in tables i and ii assumes a clean reference. if the reference noise in the bandwidth of interest is excessive, it can degrade the performance of the ad7713. a recommended reference source for the ad7713 is the ad680, a 2.5 v reference. using the ad7713 system design considerations the ad7713 operates differently from successive approxima- tion adcs or integrating adcs. since it samples the signal continuously, like a tracking adc, there is no need for a start convert command. the output register is updated at a rate determined by the first notch of the filter, and the output can be read at any time, either synchronously or asynchronously. clocking the ad7713 requires a master clock input, which may be an external ttl/cmos compatible clock signal applied to the mclk in pin with the mclk out pin left unconnected. alternatively, a crystal of the correct frequency can be connected between mclk in and mclk out, in which case the clock circuit will function as a crystal controlled oscillator. for lower clock frequencies, a ceramic resonator may be used instead of the crystal. for these lower frequency oscillators, external capacitors may be required on either the ceramic resonator or on the crystal. the input sampling frequency, the modulator sampling frequency, the ? db frequency, output update rate, and calibration time are all directly related to the master clock frequency, f clk in . reducing the master clock frequency by a factor of two will halve the above frequencies and update rate and will double the cali- bration time. the current drawn from the dv dd power supply is also directly related to f clk in . reducing f clk in by a factor of two will halve the dv dd current but will not affect the current drawn from the av dd power supply. system synchronization if multiple ad7713s are operated from a common master clock, they can be synchronized to update their output registers simul- taneously. a falling edge on the sync input resets the filter and places the ad7713 into a consistent, known state. a com- mon signal to the ad7713? sync inputs will synchronize their operation. this would normally be done after each ad7713 has performed its own calibration or has had calibration coefficients loaded to it. the sync input can also be used to reset the digital filter in systems where the turn-on time of the digital power supply (dv dd ) is very long. in such cases, the ad7713 will start operat- ing internally before the dv dd line has reached its minimum operating level, 4.75 v. with a low dv dd voltage, the ad7713? internal digital filter logic does not operate correctly. thus, the ad7713 may have clocked itself into an incorrect operating condition by the time that dv dd has reached its correct level. the digital filter will be reset upon issue of a calibration, command (w hether it is self-calibration, system calibration or background calibration) to the ad 7713. this ensures correct operation of the ad7713. in systems where the power-on default conditions of the ad7713 are acceptable, and no calibration is performed after power-on, issuing a sync pulse to the ad7713 will reset the ad7713? digital filter logic. an r, c on the sync line, w ith r, c time constant longer than the dv dd power-on time, will perform the sync function. accuracy - adcs, like vfcs and other integrating adcs, do not contain any source of nonmonotonicity, and inherently offer no missing codes performance. the ad7713 achieves excellent linearity by the use of high quality, on-chip silicon dioxide capacitors, which have a very low capacitance/voltage coefficient. the device also achieves low input drift through the use of chopper stabilized tech- niques in its input stage. to ensure excellent performance over time and temperature, the ad7713 uses digital calibration tech- niques that minimize offset and gain error. autocalibration autocalibration on the ad7713 removes offset and gain errors from the device. a calibration routine should be initi- ated on the device whenever there is a change in the ambient operating temperature or supply voltage. it should also be initiated if there is a change in the selected gain, filter notch, or bipolar/unipolar input range. however, if the ad7713 is in its background calibration mode, the above changes are all automatically taken care of (after the settling time of the filter has been allowed for). the ad7713 offers self-calibration, system calibration, and background calibration facilities. for calibration to occur on the selected channel, the on-chip microcontroller must record the modulator output for two different input conditions. these are zero-scale and full-scale points. with these readings, the microcontroller can calculate the gain slope for the input to output transfer function of the converter. internally, the part works with a resolution of 33 bits to determine its conversion result of either 16 bits or 24 bits. the ad7713 also provides the facility to write to the on-chip calibration registers, and, in this manner, the span and offset for the part can be adjusted by the user. the offset calibration register contains a value that is subtracted from all conversion results, while the full-scale calibration register contains a value that is multiplied by all conversion results. the offset calibration coefficient is sub- tracted from the result prior to the multiplication by the full-scale coefficient. in the first three modes outlined here, the drdy line indicates that calibration is complete by going low. if drdy is low before (or goes low during) the calibration command, it may take up to one modulator cycle before drdy goes high to indicate that calibration is in progress. therefore, the drdy line should be ignored for up to one modulator cycle after the last bit of the cali- bration command is written to the control register. self-calibration in the self-calibration mode with a unipolar input range, the zero-scale point used in determining the calibration coefficients is with both inputs shorted and the full-scale point is v ref . the zero-scale coefficient is determined by converting an internal shorted inputs node. the full-scale coefficient is determined from the span between this shorted inputs conversion and a conversion on an internal v ref node. the self-calibration mode
rev. d ad7713 ?7 is invoked by writing the appropriate values (0, 0, 1) to the md2, md1, and md0 bits of the control register. in this cali- bration mode, the shorted inputs node is switched in to the modulator first and a conversion is performed; the v ref node is then switched in, and another conversion is performed. when the calibration sequence is complete, the calibration coefficients updated, and the filter resettled to the analog input voltage, the drdy output goes low. the self-calibration procedure takes into account the selected gain on the pga. for bipolar input ranges in the self-calibrating mode, the sequence is very similar to that just outlined. in this case, the two points that the ad7713 calibrates are midscale (bipolar zero) and positive full scale. system calibration system calibration allows the ad7713 to compensate for system gain and offset errors as well as its own internal errors. system calibration performs the same slope factor calculations as self- calibration but uses voltage values presented by the system to the ain inputs for the zero-scale and full-scale points. system calibration is a 2-step process. the zero-scale point must be presented to the converter first. it must be applied to the converter before the calibration step is initiated and remain stable until the step is complete. system calibration is initiated by writing the appropriate values (0, 1, 0) to the md2, md1, and md0 bits of the control register. the drdy output from the device will signal when the step is complete by going low. after the zero- scale point is calibrated, the full-scale point is applied and the second step of the calibration process is initiated by again writ- ing the appropriate values (0, 1, 1) to md2, md1, and md0. again, the full-scale voltage must be set up before the calibra- tion is initiated, and it must remain stable throughout the calibration step. drdy goes low at the end of this second step to indicate that the system calibration is complete. in the unipo- lar mode, the system calibration is performed between the two endpoints of the transfer function; in the bipolar mode, it is performed between midscale and positive full scale. this 2-step system calibration mode offers another feature. after the sequence has been completed, additional offset or gain calibrations can be performed by themselves to adjust the zero reference point or the system gain. this is achieved by perform- ing the first step of the system calibration sequence (by writing 0, 1, 0 to md2, md1, and md0). this will adjust the zero- scale or offset point but will not change the slope factor from what was set during a full system calibration sequence. system calibration can also be used to remove any errors from an antialiasing filter on the analog input. a simple r, c anti- aliasing filter on the front end may introduce a gain error on the analog input voltage but the system calibration can be used to remove this error. system offset calibration system offset calibration is a variation of both the system cali- bration and self-calibration. in this case, the zero-scale point for the system is presented to the ain input of the converter. system offset calibration is initiated by writing 1, 0, 0 to md2, md1, and md0. the system zero-scale coefficient is deter- mined by converting the voltage applied to the ain input, while the full-scale coefficient is determined from the span between this ain conversion and a conversion on v ref . the zero-scale point should be applied to the ain input for the duration of the calibration sequence. this is a 1-step calibration sequence with drdy going low when the sequence is completed. in the uni- polar mode, the system offset calibration is performed between the two endpoints of the transfer function; in the bipolar mode, it is performed between midscale and positive full scale. background calibration the ad7713 also offers a background calibration mode where the part interleaves its calibration procedure with its normal conversion sequence. in the background calibration mode, the same voltages are used as the calibration points as are used in the self-calibration mode, i.e., shorted inputs and v ref . the background calibration mode is invoked by writing 1, 0, 1 to md2, md1, and md0 of the control register. w hen invoked, the background calibration mode reduces the output data rate of the ad7713 by a factor of 6 while the ? db bandwidth remains unchanged. its advantage is that the part is continually per- forming calibration and automatically updating its calibration coefficients. as a result, the effects of temperature drift, supply sensitivity and time drift on zero- and full-scale errors are auto- matically removed. when the background calibration mode is turned on, the part will remain in this mode until bits md2, md1, and md0 of the control register are changed. with back- ground calibration mode on, the first result from the ad7713 will be incorrect as the full-scale calibration will not have been performed. for a step change on the input, the second output update will have settled to 100% of the final value. table iv summarizes the calibration modes and the calibration points associated with them. it also gives the duration from when the calibration is invoked to when valid data is available to the user. span and offset limits whenever a system calibration mode is used, there are limits on the amount of offset and span that can be accommodated. the range of input span in both the unipolar and bipolar modes for ain1 and ain2 has a minimum value of 0.8 v ref /gain and a maximum value of 2.1 v ref /gain. for ain3, the mini- mum value is 3.2 v ref /gain, while the maximum value is 4.2 v ref /gain. table iv. calibration truth table zero-scale full-scale calibration type md2, md1, md0 calibration calibration sequence duration self-calibration 0, 0, 1 shorted inputs v ref 1-step 9 1/output rate system calibration 0, 1, 0 ain 2-step 4 1/output rate system calibration 0, 1, 1 ain 2-step 4 1/output rate system offset calibration 1, 0, 0 ain v ref 1-step 9 1/output rate background calibration 1, 0, 1 shorted inputs v ref 1-step 6 1/output rate
rev. d ?8 ad7713 the amount of offset that can be accommodated depends on whether the unipolar or bipolar mode is being used. this offset range is limited by the requirement that the positive full-scale calibration limit is 1.05 v ref /gain for ain1 and ain2. therefore, the offset range plus the span range cannot exceed 1.05 v ref /gain for ain1 and ain2. if the span is at its minimum (0.8 v ref /gain), the maximum the offset can be is (0.25 v ref /gain) for ain1 and ain2. for ain3, both ranges are multiplied by a factor of 4. in the bipolar mode, the system offset calibration range is again restricted by the span range. the span range of the converter in bipolar mode is equidistant around the voltage used for the zero-scale point, thus the offset range plus half the span range cannot exceed (1.05 v ref /gain) for ain1 and ain2. if the span is set to 2 v ref /gain, the offset span cannot move more than (0.05 v ref /gain) before the endpoints of the transfer function exceed the input overrange limits (1.05 v ref /gain) for ain1. if the span range is set to the minimum (0.4 v ref /gain), the maximum allowable offset range is (0.65 v ref /gain) for ain1 and ain2. the ain3 input can only be used in the unipolar mode. power-up and calibration on power-up, the ad7713 performs an internal reset, which sets the contents of the control register to a known state. how- ever, to ensure correct calibration for the device, a calibration routine should be performed after power-up. the power dissipation and temperature drift of the ad7713 are low and no warm-up time is required before the initial calibra- tion is performed. however, the external reference must have stabilized before calibration is initiated. drift considerations the ad7713 uses chopper stabilization techniques to minimize input offset drift. charge injection in the analog switches and dc leakage currents at the sampling node are the primary sources of offset voltage drift in the converter. the dc input leakage cur- rent is essentially independent of the selected gain. gain drift within the converter depends primarily upon the temperature tracking of the internal capacitors. it is not affected by leakage currents. measurement errors due to offset drift or gain drift can be elimi- nated at any time by recalibrating the converter or by operating the part in the background calibration mode. using the system calibration mode can also minimize offset and gain errors in the signal conditioning circuitry. integral and differential linearity errors are not significantly affected by temperature changes. power supplies and grounding the analog and digital supplies to the ad7713 are independent and separately pinned out to minimize coupling between the analog and digital sections of the device. the digital filter will provide rejection of broadband noise on the power supplies, except at integer multiples of the modulator sampling frequency. the digital supply (dv dd ) must not exceed the analog positive supply (av dd ) by more than 0.3 v. if separate analog and digital supplies are used, the recommended decoupling scheme is shown in figure 9. in systems where av dd = 5 v and dv dd = 5 v, it is recommended that av dd and dv dd are driven from the same 5 v supply, although each supply should be decoupled separately as shown in figure 9. it is preferable that the common supply is the system? analog 5 v supply. it is also important that power is applied to the ad7713 before signals at ref in, ain, or the logic input pins in order to avoid excessive current. if separate supplies are used for the ad7713 and the system digital circuitry, then the ad7713 should be powered up first. if it is not possible to guarantee this, then current limiting resistors should be placed in series with the logic inputs. ad7713 0.1 f 0.1 f 10 f analog supply digital 5v supply av dd dv dd figure 9. recommended decoupling scheme digital interface the ad7713? serial communications port provides a flexible arrangement to allow easy interfacing to industry-standard microprocessors, microcontrollers, and digital signal processors. a serial read to the ad7713 can access data from the output register, the control register, or from the calibration registers. a serial write to the ad7713 can write data to the control register or the calibration registers. two different modes of operation are available, optimized for different types of interface where the ad7713 can act either as master in the system (it provides the serial clock) or as slave (an external serial clock can be provided to the ad7713). these two modes, labeled self-clocking mode and external clocking mode, are discussed in detail in the following sections. self-clocking mode the ad7713 is configured for its self-clocking mode by tying the mode pin high. in this mode, the ad7713 provides the serial clock signal used for the transfer of data to and from the ad7713. this self-clocking mode can be used with processors that allow an external device to clock their serial port, including most digital signal processors and microcontrollers, such as the 68hc11 and 68hc05. it also allows easy interfacing, to serial parallel conversion circuits in systems with parallel data com- munication, allowing interfacing to 74xx299 universal shift registers without any additional decoding. in the case of shift registers, the serial clock line should have a pull-down resistor instead of the pull-up resistor shown in figure 10 and figure 11. read operation data can be read from either the output register, the control register, or the calibration registers. a0 determines whether the data read accesses data from the control register or from the output/calibration registers. this a0 signal must remain valid for the duration of the serial read operation. with a0 high, data is accessed from either the output register or from the calibration registers. with a0 low, data is accessed from the control register. the function of the drdy line is dependent on only the output update rate of the device and the reading of the output data register. drdy goes low when a new data-word is available in the output data register. it is reset high when the last bit of data (either 16th bit or 24th bit) is read from the output register. if data is not read from the output register, the drdy line will remain low. the output register will continue to be updated at the output update rate, but drdy will not indicate this. a read from the device in this circumstance will access the most recent
rev. d ad7713 ?9 word in the output register. if a new data-word becomes avail- able to the output register while data is being read from the output register, drdy will not indicate this and the new data- word will be lost to the user. drdy is not affected by reading from the control register or the calibration registers. data can be accessed from the output data register only when drdy is low. if rfs goes low with drdy high, no data trans- fer will take place. drdy does not have any effect on reading data from the control register or from the calibration registers. figure 10 shows a timing diagram for reading from the ad7713 in the self-clocking mode. this read operation shows a read from the ad7713? output data register. a read from the con- trol register or calibration registers is similar, but, in these cases, the drdy line is not related to the read function. depending on the output update rate, it can go low at any stage in the control/calibration register read cycle without affecting the read and its status should be ignored. a read operation from either the control or calibration registers must always read 24 bits of data from the respective register. figure 10 shows a read operation from the ad7713. for the timing diagram shown, it is assumed that there is a pull-up resistor on the sclk output. with drdy low, the rfs input is brought low. rfs going low enables the serial clock of the ad7713 and also places the msb of the word on the serial data line. all subsequent data bits are clocked out on a high-to-low transition of the serial clock and are valid prior to the following rising edge of this clock. the final active falling edge of sclk clocks out the lsb, and this lsb is valid prior to the final active rising edge of sclk. coincident with the next falling edge of sclk, drdy is reset high. drdy going high turns off the sclk and the sdata outputs, this means that the data hold time for the lsb is slightly shorter than for all other bits. write operation data can be written to either the control register or calibration registers. in either case, the write operation is not affected by the drdy line, and the write operation does not have any effect on the status of drdy . a write operation to the control register or the calibration register must always write 24 bits to the respective register. figure 11 shows a write operation to the ad7713. a0 determines whether a write operation transfers data to the control register or to the calibration registers. this a0 signal must remain valid for the duration of the serial write operation. the falling edge of tfs enables the internally generated sclk output. the serial data to be loaded to the ad7713 must be valid on the rising edge of this sclk signal. data is clocked into the ad7713 on the rising edge of the sclk signal, with the msb transferred first. on the last active high time of sclk, the lsb is loaded to the ad7713. subsequent to the next falling edge of sclk, the sclk output is turned off. (the timing diagram of figure 11 assumes a pull-up resistor on the sclk line.) external clocking mode the ad7713 is configured for its external clocking mode by tying the mode pin low. in this mode, sclk of the ad7713 is configured as an input, and an external serial clock must be provided to this sclk pin. this external clocking mode is designed for direct interface to systems which provide a serial clock output which is synchronized to the serial data output, including microcontrollers, such as the 80c51, 87c51, 68hc11, and 68hc05, and most digital signal processors. read operation as with the self-clocking mode, data can be read from either the output register, the control register, or the calibration registers. a0 determines whether the data read accesses data from the control register or from the output/calibration registers. this a0 three-state sdata (o) sclk (o) rfs (i) a0 (i) drdy (o) msb lsb t 3 t 2 t 4 t 5 t 6 t 9 t 10 t 8 t 7 figure 10. self-clocking mode, output data read operation sdata (i) sclk (o) tfs (i) a0 (i) msb lsb t 14 t 9 t 15 t 16 t 17 t 18 t 19 t 10 figure 11. self-clocking mode, control/calibration register write operation
rev. d ?0 ad7713 signal must remain valid for the duration of the serial read opera- tion. with a0 high, data is accessed from either the output register or from the calibration registers. with a0 low, data is accessed from the control register. the function of the drdy line is dependent on only the output update rate of the device and the reading of the output data register. drdy goes low when a new data-word is available in the output data register. it is reset high when the last bit of data (either 16th bit or 24th bit) is read from the output register. if data is not read from the output register, the drdy line will remain low. the output register will continue to be updated at the output update rate, but drdy will not indicate this. a read from the device in this circumstance will access the most recent word in the output register. if a new data-word becomes avail- able to the output register while data is being read from the output register, drdy will not indicate this, and the new data- word will be lost to the user. drdy is not affected by reading from the control register or the calibration register. data can be accessed from the output data register only when drdy is low. if rfs goes low while drdy is high, no data transfer will take place. drdy does not have any effect on read- ing data from the control register or from the calibration registers. figures 12a and 12b show timing diagrams for reading from the ad7713 in the external clocking mode. figure 12a shows a situation where all the data is read from the ad7713 in one read operation. figure 12b shows a situation where the data is read from the ad7713 over a number of read operations. both read operations show a read from the ad7713? output data register. a read from the control register or calibration registers is similar, but in these cases, the drdy line is not related to the read function. depending on the output update rate, it can go low at any stage in the control/calibration register read cycle without affecting the read and its status should be ignored. a read operation from either the control or calibration registers must always read 24 bits of data from the respective register. figure 12a shows a read operation from the ad7713 where rfs remains low for the duration of the data-word transmis- sion. with drdy low, the rfs input is brought low. the input sclk signal should be low between read and write operations. rfs going low places the msb of the word to be read on the serial data line. all subsequent data bits are clocked out on a high-to-low transition of the serial clock and are valid prior to the following rising edge of this clock. the penultimate falling edge of sclk clocks out the lsb and the final falling edge resets the drdy line high. this rising edge of drdy turns off the serial data output. figure 12b shows a timing diagram for a read operation where rfs returns high during the transmission of the word and returns low again to access the rest of the data-word. timing parameters and functions are very similar to that outlined for figure 12a, but figure 12b has a number of additional times to show timing relationships when rfs returns high in the middle of transferring a word. rfs should return high during a low time of sclk. on the rising edge of rfs , the sdata output is turned off. drdy remains low and will remain low until all bits of the data-word are read from the ad7713, regardless of the number of times rfs changes state during the read operation. depending on the time between the falling edge of sclk and the rising edge of rfs , the next bit (bit n + 1) may appear on the data bus before rfs goes high. when rfs returns low again, it activates the sdata output. when the entire word is transmitted, the sdata (o) sclk (i) three-state rfs (i) a0 (i) t 22 msb lsb drdy (o) t 21 t 20 t 23 t 26 t 29 t 28 t 27 t 25 t 24 figure 12a. external clocking mode, output data read operation three-state msb bit n bit n+1 sdata (o) sclk (i) rfs (i) a0 (i) drdy (o) t 22 t 20 t 31 t 26 t 24 t 25 t 27 t 30 t 24 t 25 figure 12b. external clocking mode, output data read ( rfs returns high during read operation)
rev. d ad7713 ?1 drdy line will go high, turning off the sdata output as per figure 12a. write operation data can be written to either the control register or calibration registers. in either case, the write operation is not affected by the drdy line, and the write operation does not have any effect on the status of drdy . a write operation to the control register or the calibration register must always write 24 bits to the respective register. figure 13a shows a write operation to the ad7713 with tfs remaining low for the duration of the write operation. a0 determines whether a write operation transfers data to the con- trol register or to the calibration registers. this a0 signal must remain valid for the duration of the serial write operation. as before, the serial clock line should be low between read and write operations. the serial data to be loaded to the ad7713 must be valid on the high level of the externally applied sclk signal. data is clocked into the ad7713 on the high level of this sclk signal with the msb transferred first. on the last active high time of sclk, the lsb is loaded to the ad7713. figure 13b shows a timing diagram for a write operation to the ad7713 with tfs returning high during the write operation and returning low again to write the rest of the data-word. tim- ing parameters and functions are very similar to that outlined for figure 13a, but figure 13b has a number of additional times to show timing relationships when tfs returns high in the middle of transferring a word. data to be loaded to the ad7713 must be valid prior to the rising edge of the sclk signal. tfs should return high during the low time of sclk. after tfs returns low again, the next bit of the data-word to be loaded to the ad7713 is clocked in on the next high level of the sclk input. on the last active high time of the sclk input, the lsb is loaded to the ad7713. simplifying the external clocking mode interface in many applications, the user may not require the facility of writing to the on-chip calibration registers. in this case, the serial interface to the ad7713 in external clocking mode can be simplified by connecting the tfs line to the a0 input of the ad 7713 (see figure 14). th is means that any write to the device will load data to the control register (since a0 is low while tfs is low), and any read to the device will access data from the output data register or from the calibration registers (since a0 is high while rfs is l ow) . it should be noted that in this arrangement, the user does not have the capability of read- ing from the control register. another method of simplifying the interface is to generate the tfs signal from an inverted rfs signal. however, generating the signals the opposite way around ( rfs from an inverted tfs ) will cause writing errors. four interface lines rfs ad7713 sdata sclk tfs a0 figure 14. simplified interface with tfs connected to a0 sclk (i) sdata (i) tfs (i) a0 (i) msb lsb t 32 t 33 t 34 t 26 t 27 t 36 t 35 figure 13a. external clocking mode, control/calibration register write operation sclk (i) sdata (i) tfs (i) a0 (i) msb bit n bit n+1 t 32 t 26 t 30 t 27 t 36 t 35 t 35 t 36 figure 13b. external clocking mode, control/calibration register write operation ( tfs returns high during write operation)
rev. d ?2 ad7713 microcomputer/microprocessor interfacing the ad7713? flexible serial interface allows easy interface to most microcomputers and microprocessors. figure 15 shows a flowchart diagram for a typical programming sequence for read- ing data from the ad7713 to a microcomputer, while figure 16 shows a flowchart diagram for writing data to the ad7713. figures 17 and 18 show some typical interface circuits. the flowchart in figure 15 is for continuous read operations from the ad7713 output register. in the example shown, the drdy line is continuously polled. depending on the micropro- cessor configuration, the drdy line may come to an interrupt input, in which case the drdy will automatically generate an interrupt without being polled. reading the serial buffer could be anything from one read operation up to three read operations (where 24 bits of data are read into an 8-bit serial register). a read operation to the control/calibration registers is similar, but, in this case, the status of drdy can be ignored. the a0 line is brought low when the rfs line is brought low when reading from the control register. configure and initialize c/ p serial port bring rfs low poll drdy bring rfs , tfs and high start 3 bring rfs high read serial buffer reverse order of bits drdy low? yes no figure 15. flowchart for continuous read operation to the ad7713 the flowchart also shows the bits being reversed after they have been read in from the serial port. this depends on whether the microprocessor expects the msb of the word first or the lsb of the word first. the ad7713 outputs the msb first. the flowchart in figure 16 is for a single 24-bit write operation to the ad7713 control or calibration registers. this shows data being transferred from data memory to the accumulator before being written to the serial buffer. some microprocessor systems will allow data to be written directly to the serial buffer from data memory. writing data to the serial buffer from the accumulator will generally consist of either two or three write operations, depending on the size of the serial buffer. the flowchart also shows the option of the bits being reversed before being written to the serial buffer. this depends on whether the first bit transmitted by the microprocessor is the msb or the lsb. the ad7713 expects the msb as the first bit in the data stream. in cases where the data is being read or being written in bytes and the data has to be reversed, the bits will have to be reversed for every byte. configure and initialize c/ p serial port bring rfs and a0 low bring rfs , tfs and a0 high start end 3 load data from address to accumulator write data from accumulator to serial buffer bring tfs and a0 high reverse order of bits figure 16. flowchart for single write operation to the ad7713 ad7713 to 8xc51 interface figure 17 shows an interface between the ad7713 and the 8xc51 microcontroller. the ad7713 is configured for its external clock- ing mode, while the 8xc51 is configured in its mode 0 serial interface mode. the drdy line from the ad7713 is connected to the port p1.2 input of the 8xc51, so the drdy line is polled by the 8xc51. the drdy line can be connected to the int1 input of the 8xc51 if an interrupt driven system is preferred. rfs ad7713 sdata sclk tfs a0 p1.0 p3.0 p3.1 p1.1 p1.2 mode drdy p1.3 sync dv dd 8xc51 figure 17. ad7713 to 8xc51 interface
rev. d ad7713 ?3 table v shows some typical 8xc51 code used for a single 24-bit read from the output register of the ad7713. table v shows some typical code for a single write operation to the control register of the ad7713. the 8xc51 outputs the lsb first in a write operation while the ad7713 expects the msb first, so the data to be transmitted has to be rearranged before being written to the output serial register. similarly, the ad7713 outputs the msb first during a read operation while the 8xc51 expects the lsb first. therefore, the data which is read into the serial buffer needs to be rearranged before the correct data-word from the ad7713 is available in the accumulator. table v. 8xc51 code for reading from the ad7713 mov scon,#00010001b; configure 8051 for mode 0 mov ie,#00010000b; disable all interrupts setb 90h; set p1.0, used as rfs setb 91h; set p1.1, used as tfs setb 93h; set p1.3, used as a0 mov r1,#003h; sets number of bytes to be read in a read operation mov r0,#030h; start address for where bytes will be loaded mov r6,#004h; use p1.2 as drdy wait: nop; mov a,p1; read port 1 anl a,r6; mask out all bits except drdy jz read; if zero read sjmp wait; otherwise keep polling read: clr 90h; bring rfs low clr 98h; clear receive flag poll: jb 98h, read1 tests receive interrupt flag sjmp poll read 1: mov a,sbuf; read buffer rlc a; rearrange data mov b.0,c; reverse order of bits rlc a; mov b.1,c; rlc a; mov b.2,c; rlc a; mov b.3,c; rlc a; mov b.4,c; rlc a; mov b.5,c; rlc a; mov b.6,c; rlc a; mov b.7,c; mov a,b; mov @r0,a; write data to memory inc r0; increment memory location dec r1 decrement byte counter mov a,r1 jz end jump if zero jmp wait fetch next byte end: setb 90h bring rfs h igh fin: sjmp fin table vi. 8xc51 code for writing to the ad7713 mov scon,#00000000b; configure 8051 for mode 0 operation and enable serial reception mov ie,#10010000b; enable transmit interrupt mov ip,#00010000b; prioritize the transmit interrupt setb 91h; bring tfs h igh setb 90h; bring rfs h igh mov r1,#003h; sets number of bytes to be written in a write operation mov r0,#030h; start address in ram for bytes mov a,#00h; clear accumulator mov sbuf,a; initialize the serial port wait: jmp wait; wait for interrupt int routine: nop; interrupt subroutine mov a,r1; load r1 to accumulator jz fin; if zero jump to fin dec r1; decrement r1 byte counter mov a,@r; move byte into the accumulator inc r0; increment address rlc a; rearrange data?rom lsb first to msb first mov b.0,c; rlc a; mov b.1,c; rlc a; mov b.2,c; rlc a; mov b.3,c; rlc a; mov b.4,c; rlc a; mov b.5,c; rlc a; mov b.6,c; rlc a: mov b.7,c; mov a,b; clr 93h; bring a0 low clr 91h; bring tfs low mov sbuf,a; write to serial port reti; return from subroutine fin: setb 91h; set tfs h igh setb 93h; set a0 high reti; return from interrupt subroutine ad7713 to 68hc11 interface figure 18 shows an interface between the ad7713 and the 68hc11 microcontroller. the ad7713 is configured for its exter- nal clocking mode, while the spi port is used on the 68hc11, which is in its single chip mode. the drdy line from the ad7713 is connected to the port pc2 input of the 68hc11, so the drdy line is polled by the 68hc11. the drdy line can be connected to the irq input of the 68hc11 if an interrupt driven system is preferred. the 68hc11 mosi and miso lines should be config- ured for wired-or operation. depending on the interface configuration, it may be necessary to provide bidirectional buffers between the 68hc11 mosi and miso lines. the 68hc11 is configured in the master mode with its cpol bit set to a logic 0 and its cpha bit set to a logic 1.
rev. d ?4 ad7713 sync rfs tfs a0 sdata sclk mode pc0 pc1 pc2 pc3 miso mosi 68hc11 ad7713 drdy ss sck dv dd dv dd figure 18. ad7713 to 68hc11 interface applications 4-wire rtd configurations figure 19 shows a 4-wire rtd application where the rtd transducer is interfaced directly to the ad7713. in the 4-wire configuration, there are no errors associated with lead resis- tances as no current flows in the measurement leads connected to ain1(+) and ain1(?. one of the rtd current sources is used to provide the excitation current for the rtd. a common nominal resistance value for the rtd is 100 ? and, therefore, the rtd will generate a 20 mv signal, which can be handled directly by the analog input of the ad7713. in the circuit shown, the second rtd excitation current is used to generate the reference voltage for the ad7713. this reference voltage is developed across r ref and applied to the differential reference inputs. for the nominal reference voltage of 2.5 v, r ref is 12.5 k ? . this scheme ensures that the analog input voltage span remains ratiometric to the reference voltage. any errors in the analog input voltage due to the temperature drift of the rtd current source is compensated for by the variation in the reference volt- age. the typical matching between the two rtd current sources is less than 3 ppm/ c. ain1(+) ain1(?) av dd agnd dgnd a = 1 ? 128 ad7713 rtd1 ref in(+) ref in(?) rtd2 200 a 5v r ref internal circuitry dv dd pga 200 a rtd figure 19. 4-wire rtd application with the ad7713 3-wire rtd configurations figure 20 shows a 3-wire rtd configuration using the ad7713. in the 3-wire configuration, the lead resistances will result in errors if only one current source is used as the 200 a will flow through rl1 developing a voltage error between ain1(+) and ain1(?. in the scheme outlined below, the second rtd cur- rent source is used to compensate for the error introduced by the 200 a flowing through rl 1. the second rtd current flows through rl2. assuming rl1 and rl2 are equal (the leads would normally be of the same material and of equal length) and rtd1 and rtd2 match, then the error voltage across rl2 equals the error voltage across rl1, and no error voltage is developed between ain1(+) and ain1(?. twice the voltage is developed across rl3 but since this is a common-mode voltage, it will not introduce any errors. the reference voltage is derived from one of the current sources. this gives all the benefits of eliminating rtd tem perature coefficient errors as outlined in figure 19. the voltage on either rtd input can go to within 2 v of the av dd supply. the circuit is shown for a 2.5 v reference. ain(+) ain(?) av dd dv dd agnd dgnd a = 1 ? 128 ad7713 rtd1 rtd2 12.5k internal circuitry ref in(+) ref in(?) r l1 r l2 r l3 pga rtd 200 a 200 a figure 20. 3-wire rtd application with the ad7713 4?0 ma loop the ad7713? high level input can be used to measure the current in 4?0 ma loop applications as shown in figure 21. in this case, the system calibration capabilities of the ad7713 can be used to remove the offset caused by the 4 ma flowing through the 500 ? resistor. the ad7713 can handle an input span as low as 3.2 v ref (= 8 v with a v ref of 2.5 v) even though the nominal input voltage range for the input is 10 v. therefore, the full span of the adc can be used for measuring the current between 4 and 20 ma. ref in(+) ain1(+) ain1(?) ain3 agnd dgnd a = 1 ? 128 ref in(?) ad7713 4?20ma loop analog 5v supply 500 voltage attenuation av dd av dd dv dd internal circuitry mux pga 1 a figure 21. 4-20 ma measurement using the ad7713
rev. d ad7713 ?5 other 24-bit signal conditioning adcs available from analog devices ad7710 features charge balancing 24 bits no missing codes 0.0015% nonlinearity 2-channel programmable gain front end gains from 1 to 128 differential inputs low-pass filter with programmable filter cutoffs ability to read/write calibration coefficients bidirectional microcontroller serial interface internal/external reference option single- or dual-supply operation low power (25 mw typ) with power-down mode (7 mw typ) applications weigh scales thermocouples smart transmitters chromatography ad7711 features charge balancing adc 24 bits no missing codes 0.0015% nonlinearity 2-channel programmable gain front end gains from 1 to 128 1 differential input 1 single-ended input low-pass filter with programmable filter cutoff ability to read/write calibration coefficients rtd excitation current sources bidirectional microcontroller serial interface internal/external reference option single- or dual-supply operation low power (25 mw typ) with power-down mode (7 mw typ) applications rtd transducers process control smart transmitters portable industrial instruments functional block diagram charging balancing a/d converter auto-zeroed ? modulator digital filter clock generation ad7710 mux agnd dgnd mclk out mclk in ain1(+) ain1(?) ain2(+) ain2(?) ref in(+) av dd dv dd av dd 4.5 a a = 1 ? 128 v ss av dd 20 a v bias ref out ref in(?) sync rtd current 2.5v reference pga serial interface control register output register mode sdata a0 drdy tfs rfs sclk functional block diagram charging balancing a/d converter auto-zeroed ? modulator digital filter clock generation ad7711 mux agnd dgnd mclk out mclk in ain1(+) ain1(?) ain2 ref in(+) av dd dv dd av dd 4.5 a a = 1 ? 128 v ss v bias ref out ref in(?) sync rtd2 rtd1 2.5v reference pga av dd 200 a 200 a serial interface control register output register mode sdata a0 drdy tfs rfs sclk
rev. d ?6 ad7713 ad7712 features charge balancing adc 24 bits no missing codes 0.0015% nonlinearity high level and low level analog input channels programmable gain for both inputs gains from 1 to 128 differential input for low level channel low-pass filter with programmable filter cutoffs ability to read/write calibration coefficients bidirectional microcontroller serial interface internal/external reference option single- or dual-supply operation low power (25 mw typ) with power-down mode (100 w typ) applications process control smart transmitters portable industrial instruments functional block diagram charging balancing a/d converter auto-zeroed ? modulator digital filter clock generation ad7712 mux agnd dgnd mclk out mclk in ain1(+) ain1(?) ref in(+) av dd dv dd av dd 4.5 a a = 1 ? 128 v ss v bias ref out ref in(?) sync standb y tp ain2 2.5v reference pga serial interface control register output register mode sdata a0 drdy tfs rfs sclk voltage attenuation
rev. d ad7713 ?7 outline dimensions 24-lead plastic dual in-line package [pdip] (n-24) dimensions shown in inches and (millimeters) 24 1 12 13 1.185 (30.01) 1.165 (29.59) 1.145 (29.08) 0.295 (7.49) 0.285 (7.24) 0.275 (6.99) 0.150 (3.81) 0.135 (3.43) 0.120 (3.05) 0.015 (0.38) 0.010 (0.25) 0.008 (0.20) 0.325 (8.26) 0.310 (7.87) 0.300 (7.62) seating plane 0.015 (0.38) min 0.180 (4.57) max 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 0.150 (3.81) 0.130 (3.30) 0.110 (2.79) 0.100 (2.54) bsc 0.060 (1.52) 0.050 (1.27) 0.045 (1.14) controlling dimensions are in inches; millimeter dimensions (in parentheses) are rounded-off inch equivalents for reference only and are not appropriate for use in design compliant to jedec standards mo-095ag 24-lead standard small outline package [soic] wide body (rw-24) dimensions shown in millimeters and (inches) controlling dimensions are in millimeters; inch dimensions (in parentheses) are rounded-off millimeter equivalents for reference only and are not appropriate for use in design compliant to jedec standards ms-013ad 8 0 0.75 (0.0295) 0.25 (0.0098) 45 1.27 (0.0500) 0.40 (0.0157) seating plane 0.30 (0.0118) 0.10 (0.0039) 2.65 (0.1043) 2.35 (0.0925) 1.27 (0.0500) bsc 24 13 12 1 10.65 (0.4193) 10.00 (0.3937) 7.60 (0.2992) 7.40 (0.2913) 15.60 (0.6142) 15.20 (0.5984) coplanarity 0.10 0.33 (0.0130) 0.20 (0.0079) 0.51 (0.0201) 0.31 (0.0122) 24-lead ceramic dual in-line package [cerdip] (q-24) dimensions shown in inches and (millimeters) 24 112 13 0.310 (7.87) 0.220 (5.59) pin 1 0.005 (0.13) min 0.098 (2.49) max 15 0 0.320 (8.13) 0.290 (7.37) 0.015 (0.38) 0.008 (0.20) seating plane 0.200 (5.08) max 1.280 (32.51) max 0.150 (3.81) min 0.200 (5.08) 0.125 (3.18) 0.023 (0.58) 0.014 (0.36) 0.100 (2.54) bsc 0.070 (1.78) 0.030 (0.76) 0.060 (1.52) 0.015 (0.38) controlling dimensions are in inches; millimeter dimensions (in parentheses) are rounded-off inch equivalents for reference only and are not appropriate for use in design
rev. d c01553??/04(d) ?8 ad7713 revision history location page 3/04?ata sheet changed from rev. c to rev. d. updated layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . universal changes to functional block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 changes to specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 changes to ordering guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 changes to self-calibration section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 changes to ad7713 to 68hc11 interface section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 deleted ad7713 to adsp-2105 interface section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 deleted figure 19 and renumbered succeeding figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 updated outline dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

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