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hd74ac series common information september 2000 customer service division semiconductor & integrated circuits hitachi, ltd. disclaimer 1. hitachi neither warrants nor grants licenses of any rights of hitachi? or any third party? patent, copyright, trademark, or other intellectual property rights for information contained in this document. hitachi bears no responsibility for problems that may arise with third party? rights, including intellectual property rights, in connection with use of the information contained in this document. 2. products and product specifications may be subject to change without notice. confirm that you have received the latest product standards or specifications before final design, purchase or use. 3. hitachi makes every attempt to ensure that its products are of high quality and reliability. however, contact hitachi? sales office before using the product in an application that demands especially high quality and reliability or where its failure or malfunction may directly threaten human life or cause risk of bodily injury, such as aerospace, aeronautics, nuclear power, combustion control, transportation, traffic, safety equipment or medical equipment for life support. 4. design your application so that the product is used within the ranges guaranteed by hitachi particularly for maximum rating, operating supply voltage range, heat radiation characteristics, installation conditions and other characteristics. hitachi bears no responsibility for failure or damage when used beyond the guaranteed ranges. even within the guaranteed ranges, consider normally foreseeable failure rates or failure modes in semiconductor devices and employ systemic measures such as fail- safes, so that the equipment incorporating hitachi product does not cause bodily injury, fire or other consequential damage due to operation of the hitachi product. 5. this product is not designed to be radiation resistant. 6. no one is permitted to reproduce or duplicate, in any form, the whole or part of this document without written approval from hitachi. 7. contact hitachi? sales office for any questions regarding this document or hitachi semiconductor products.
fact descriptions and family characteristics 1 fact descriptions and family characteristics 1. advanced cmos fact logic fact logic offers a unique combination of high speed, low power dissipation, high noise immunity, wide fanout capability, extended power supply range and high reliability. this data book describes the product line with device specifications as well as material discussing design considerations and comparing the fact family to predecessor technologies. for direct replacement of ls, als and other ttl devices, hd74act circuits with ttl-type input thresholds are included in the fact family. these include the more popular bus drivers/transceivers as well as many other hd74actxxx devices. 1.1 characteristics full logic product line industry standard functions and pinouts for ssi and msi meets or exceeds jedec standards for hd74acxx family ttl inputs on selected circuits high performance outputs ? common output structure for standard and buffer drivers ? output sink/source current of 24 ma ? transmission line driving 50 w guaranteed operation from 2 to 6 volts guaranteed temperature range e40 c to +85 c improved esd protection network high current latch-up immunity 1.2 interfacing fact devices have a wide operating voltage range (v cc = 2 to 6 vdc) and sufficient current drive to interface with most other logic families available today. device designators are as follows: hd74ac is a high-speed cmos device with cmos input switching levels and buffered cmos outputs that can drive 24 ma of i oh and i ol current. industry standard hd74ac nomenclature and pinouts are used. hd74act is a high-speed cmos device with a ttl-to-cmos input buffer stage. these device inputs are designed to interface with ttl outputs operating with a v cc = 5 v 0.5 v with v oh = 2.4 v and v ol = 0.4 v, but are functional over the entire fact operating voltage range of 2.0 to 5.5 vdc. these devices have buffered outputs that will drive cmos or ttl devices with no additional interface circuitry. hd74act devices have the same output structures as hd74ac devices.
fact descriptions and family characteristics 2 1.3 low power cmos operation if there is one single characteristic that justifies the existence of cmos, it is low power dissipation. in the quiescent state, fact draws three orders of magnitude less power than the equivalent ls or als ttl device. this enhances system reliability; because costly regulated high current power supplies, heat sinks and fans are eliminated. fact logic devices are ideal for portable systems such as laptop computers and backpack communications systems. operating power is also very low for fact logic. power consumption of various technologies with a clock frequency of 1 mhz is shown below. fact = 0.1 mw/gate als = 1.2 mw/gate ls = 2.0 mw/gate hc = 0.1 mw/gate figure 1 illustrates the effects of i cc versus power supply voltage (v cc ) for two load capacitance values: 50 pf and stray capacitance. the clock frequency was 1 mhz for the measurements. c l = 50 pf @ 1 mhz i cc ( m a) c l = fixture capacitance @ 1 mhz 6 4 v cc (volts) 2 0 0 100 200 300 400 500 figure 1 i cc vs v cc 1.4 ac performance in comparison to ls, als and hc families, fact devices have faster internal gate delays as well as the basic gate delays. additionally, as the level of integration increases, fact logic leads the way to very high-speed systems. the example below describes typical values for a hd74xx138, 3 to 8 line decoder. fact = 6.0 ns @ c l = 50 pf als = 12.0 ns @ c l = 50 pf ls = 27.0 ns @ c l = 15 pf hc = 17.0 ns @ c l = 50 pf
fact descriptions and family characteristics 3 ac performance specifications are guaranteed at 5.0 v 0.5 v and 3.3 v 0.3 v. for worst case design at 2.0 v v cc on all device types, the formula below can be used to determine ac performance. ac performance at 2.0 v v cc = 1.9 ac specification at 3.3 v 1.5 multiple output switching propagation delay is affected by the number of outputs switching simultaneously. typically, devices with more than one output will follow the rule: for each output switching, derate the databook specification by 250 ps. this effect typically is not significant on an octal device unless more than four outputs are switching simultaneously. this derating is valid for the entire temperature range and 5.0 v 10% v cc . 1.6 noise immunity the noise immunity of a logic family is also an important equipment cost factor in terms of decoupling components, power supply dynamic resistance and regulation as well as layout rules for pc boards and signal cables. the comparisons shown describe the difference between the input threshold of a device and the output voltage, |v il ? ol | / |v ih ? oh | at 4.5 v v cc . fact = 1.25/1.25 v als = 0.4/0.7 v ls = 0.3/0.7 v@ 4.75 v v cc hc = 1.25/1.25 v 1.7 output characteristics all fact outputs are buffered to ensure consistent output voltage and current specifications across the family. both hd74ac and hd74act device types have the same output structures. two clamp diodes are internally connected to the output pin to suppress voltage overshoot and undershoot in noisy system applications witch can result from impedance mismatching. the balanced output design allows for controlled edge rates and equal rise and fall times. all devices (hd74ac or hd74act) are guaranteed to source and sink 24 ma. hd74ac/actxxx, are capable of driving 50 w transmission lines. 1.7.1 i ol /i oh characteristics fact = 24/e24 ma als = 24/e15 ma ls = 8/e0.4 ma @ 4.75 v v cc hc = 4/e4 ma
fact descriptions and family characteristics 4 1.7.2 dynamic output drive traditionally, in order to predict what incident wave voltages would occur in a system, the designer was required to do an output analysis using a bergeron diagram. not only is this a long and time-consuming operation, but the designer needed to depend upon the accuracy and reliability of the manufacturer-supplied ?ypical?output i/v curve. additionally, there was no way to guarantee that any supplied device would meet these ?ypical?performance values across the operating voltage and temperature limits. hitachi has taken the necessary steps to guarantee incident wave switching on transmission lines with impedances as low as 50 w . figure 2 shows a bergerton diagram for switching both high-to-low and low-to-high. on the right side of the graph (i out > 0), are the v oh and i ih curves for fact logic while on the left side (i out < 0), are the curves for v ol and i il . although we will only discuss here the low-to-high transition, the information presented may be applied to a high-to-low transition. 0 current (a) 0.1 0.2 ?.1 ?.2 ? ? 0 1 2 3 4 5 6 7 volts (v) v oh /i oh high-to-low low-to-high line 1 slope = 50 w line 2 slope = 50 w v in /i in v ol /i ol figure 2 gate driving 50 w line reflection diagram begin analysis at the v ol (quiescent) point. this is the intersection of the v ol /i ol curve for the output and the v in /i in curve for the input. for cmos inputs and outputs, this point will be approximately 100 mv. then draw a 50 w load line from this intersection to the v oh /i oh curve as shown by line 1. this intersection is the voltage that the incident wave will have. here it occurs at approximately 3.95 v. then draw a line with a slope of e50 w from this first intersection point to the v in /i in curve as shown by line 2. this second intersection will be the first reflection back from the input gate. continue this process of drawing the load line from each intersection to the next. lines terminating on the v oh /i oh curve should have positive slopes while lines terminating on the v in /i in curve should have negative slopes. each intersection point predicts the voltage of each reflected wave on the transmission line. intersection points on the v oh /i oh curve will be waves travelling from the driver to the receiver while intersection points on the v in /i in curve will be waves travelling from the receiver to the driver. figure 3 and 4 show the resultant waveforms. each division on the time scale represents the propagation delay of the transmission line.
fact descriptions and family characteristics 5 while this exercise can be done for fact, it is no longer necessary. fact is guaranteed to drive an incident wave of enough voltage to switch another fact input. volts (v) time (1 propagation delay/div) 0123456789 receiver driver 10 11 ? ? 0 1 2 3 4 5 6 7 receiver driver volts (v) time (1 propagation delay/div) 01234567891011 ? ? 0 1 2 3 4 5 6 7 figure 3 resultant waveforms driving 50 w line?heoretical receiver driver vzr 0 volts (v) time (ns) 50 100 150 200 250 300 350 400 450 ? 0 1 2 3 4 5 6 7 receiver vzr 0 driver volts (v) time (ns) 50 100 150 200 250 300 350 400 450 ? 0 1 2 3 4 5 6 7 figure 4 resultant waveforms driving 50 w line?ctual
fact descriptions and family characteristics 6 we can calculate what current is required by looking at the bergeron diagram. the quiescent voltage on the line will be within 100 mv of either rail. we know what voltage is required to guarantee a valid voltage at the receiver. this is either 70% or 30% of v cc . the formula for calculating the current and voltage required is | (v oq ? i )/z o | at v i . for v oq = 100 mv, v ih = 3.85 v, v cc = 5.5 v and z o = 50 w , the required i oh at 3.85 v is 75 ma. for the high-to-low transition, v qq = 5.4 v, v il = 1.65 v, and z o = 50 w , i ol is 75 ma at 1.65 v. fact?s i/o specifications include these limits. for transmission lines with impedances greater than 50 w , the current requirements are less and switching is still guaranteed. it is important to note that the typical 24 ma drive specification is not adequate to guarantee incident wave switching. the only way to guarantee this is to guarantee the current required to switch a transmission line from the output quiescent point to the valid v in level. the performance charts in figures 5 to 7 are provided in order to aid the designer in determining dynamic output current drive of fact devices with various power supply voltage. voltage (v) current (ma) ?0 ?00 ?50 ?00 ma 0 ? 2 ? 1 0 3 4 5 6 7 v cc = 4.5 v v cc = 5.5 v v cc = 5.0 v figure 5 output characteristics v oh /i oh , hd74ac00 current (ma) 150 100 50 200 ma 0 voltage (v) ? 2 ? 1 0 3 4 5 6 7 v cc = 4.5 v v cc = 5.0 v v cc = 5.5 v figure 6 output characteristics v ol /i ol , hd74ac00
fact descriptions and family characteristics 7 voltage (v) current (ma) ?.1 0 0.1 v in /i in 0.2 ?.2 ? 2 ? 1 0 3 4 5 6 7 v cc = 5.0 v, ta = 25 c figure 7 input characteristics v in /i in 1.8 choice of voltage specifications to obtain better performance and higher density, semiconductor technologies are reducing the vertical and horizontal dimensions of integrated device structures. due to a number of electrical limitations in the manufacture of vlsi devices and the need for low-voltage operation in memory cards, the jedec committee decided to establish interface standards for devices operating at 3.3 v 0.3 v. to this end, hitachi guarantees all of its devices operational at 3.3 v 0.3 v. note also that ac and dc specifications are guaranteed between 3.0 and 5.5 v. operation of fact logic is also guaranteed from 2.0 to 6.0 v on v cc . operating voltage ranges fact = 2.0 to 6.0 v als = 5.0 v 10% ls = 5.0 v 5% hc = 2.0 to 6.0 v 1.9 fact replaces ls, als, hcmos the advanced cmos family is specifically designed to outperform the ls, als, and hcmos families. figure 8 shows the relative position of various logic families in speed/power performance. table 1 summarizes the key performance specifications for various competitive technology logic families. table 2 compares propagation delays.
fact descriptions and family characteristics 8 0.001 0.3 1.0 3.0 10.0 2 4 6 8 10 hc fact als ls fast as propagation delay (ns) power per gate (mw) ( not to scale ) figure8 propagation delays table 1a logic family comparisons general characteristics (all max ratings) fact characteristics symbol ls als hcmos hd74ac hd74act unit operating voltage range v cc/ee/dd 5 5% 5 10% 2.0 to 6.0 2.0 to 6.0 4.5 to 5.5 v operating temperature range ?0 to +75 0 to +70 ?0 to +85 ?0 to +85 ?0 to +85 c input voltage (limits) v ih (min) 2.0 2.0 3.15 3.15 2.0 v v il (max) 0.8 0.8 1.35 1.35 0.8 v output voltage (limits) v oh (min) 2.7 2.7 v cc ?0.1 v cc ?0.1 v cc ?0.1 v v ol (max) 0.5 0.5 0.1 0.1 0.1 v input current i ih 20 20 +1.0 +1.0 +1.0 m a i il ?00 ?00 ?.0 ?.0 ?.0 m a output current at v 0 (limit) i oh ?.4 ?.4 ?.0 @ v cc 0.37 ?4 @ v cc 0.7 ?4 @ v cc 0.7 ma i ol 8.0 8.0 4.0 @ 0.33 v 24 @ 0.37 v 24 @ 0.37 v ma dc noise margin low/high dcm 0.3/0.7 0.4/0.7 1.25/1.25 1.25/1.25 0.7/2.4 v
fact descriptions and family characteristics 9 table 1b logic family comparisons speed/power characteristics (all typical ratings) fact characteristics symbol ls als hcmos hd74ac hd74act unit quiescent supply current/gate i g 0.4 0.2 0.0005 0.0005 ma power/gate (quiescent) p g 2.0 1.2 0.0025 0.0025 mw propagation delay t p 10 5.0 8.0 5.0 ns speed power product 20 6.0 0.02 0.01 pj clock frequency d/ff f max 33 50 50 160 mhz table 2 propagation delay product ls als hcmos fact unit t plh /t phl hd74xx00 typ 10.0 5.0 8.0 5.0 ns max 15.0 11.0 23.0 8.5 ns t plh /t phl (clock to q) hd74xx74 typ 25.0 12.0 14.0 8.0 ns max 40.0 18.0 40.0 10.5 ns t plh /t phl (clock to q) hd74xx163 typ 18.0 10.0 18.0 5.0 ns max 27.0 17.0 40.0 10.0 ns note: continuous: (ls) v cc = 5.0 v, c l = 15pf, 25 c; (als/hc/fact) v cc = 5.0 v 10%, c l = 50 pf, typ values at 25?c, max values at 0 to 70?c for als. ?0 to +85?c for hc/fact.
fact descriptions and family characteristics 10 2. circuit characteristics 2.1 power dissipation one advantage to using cmos logic is its extremely low power consumption. during quiescent conditions, fact will consume several orders of magnitude less current than its bipolar counterparts. but dc power consumption is not the whole picture. any circuit will have ac power consumption, whether it is built with cmos or bipolar technologies. power consumption of a circuit can be calculated using the formula: p d = [(c l + c pd ) ?v cc ?v s ?f] + [i q ?v cc ] where: p d = power dissipation c l = load capacitance c pd = device power capacitance v cc = power supply v s = output voltage swing f = frequency of operation i q = quiescent current power consumption for fact is dependent on the supply voltage, frequency of operation, internal capacitance and load. v s will be v cc and i q can be considered negligible for cmos. therefore, the simplified formula for cmos is: p d = (c l + c pd )v cc 2 f c pd values for cmos devices are calculated by measuring the power consumption of a device at two different frequencies. c pd is calculated in the following manner. 1. the power supply voltage is set to v cc = 5.0 vdc. 2. signal inputs are set up so that as many outputs as possible are switching, giving a worst-case situation per jedec c pd conditions.
fact descriptions and family characteristics 11 v cc set clr d clk q q ?4 set clr d clk q q ?4 set clr d clk q q ?4 set clr d clk q q ?4 e1 e2 e3 a0 a1 a2 01 02 03 04 05 06 07 08 input ?4 load devices ?38 figure 9 power demonstration circuit schematic 3. the power supply current is measure and recorded at input frequencies of 200 khz and 1 mhz. 4. the power dissipation capacitance is calculated by solving the two simultaneous equations p 1 = (c pd ?v cc 2 ?f 1 ) + (i cc ?v cc ) p 2 = (c pd ?v cc 2 ?f 2 ) + (i cc ?v cc ) giving c pd = (p 1 ?p 2 )/v cc 2 (f 1 ?f 2 ) or c pd = (i 1 ?i 2 )/v cc (f 1 ?f 2 ) where i 1 = supply current at f 1 = 200 khz. i 2 = supply current at f 2 = 1 mhz
fact descriptions and family characteristics 12 on fact device data sheets, c pd is a typical value and is given either for the package or for the individual device (i.e., gates, flip-flops, etc.) within the package. the circuit shown in figure 9 was used to compare the power consumption of fact versus als devices. two identical circuits were built on the same board and driven from the same input. in the circuit, the input signal was driven into four d-type flip-flops which act as divide-by-2 frequency dividers. the outputs from the flip-flops were connected to the inputs of a hd74ac138/74als138 decoder. this generated eight non-overlapping clock pulses on the outputs of the hd74ac138/74als138, which were then connected to an hd74ac04/74als04 inverter. the input frequency was then varied and the power consumption was measured. figure 10 illustrates the results of these measurements. power (mw) 60 80 100 40 als als circuit stopped functioning fact frequency (mhz) 20 0 0 50 100 200 150 300 250 350 400 figure 10 fact vs als circuit power below 40 mhz, the fact circuit dissipates much less power than the als version. it is interesting to note that when the frequency went to zero, the fact circuit? power consumption also went to zero; the als circuit continued to dissipate almost 100 mw. another advantage of fact is its capabilities above 40 mhz. at this frequency, the first 74als74 d-type flip-flop ceased to operate. once this occurred, the entire circuit stopped working and the power consumption fell to its quiescent value. the fact device, however, continued functioning beyond the limit of the frequency generator, which was 100 mhz. this graph shows two advantages of fact circuits (power and speed). fact logic delivers increased performance in addition to offering the power savings of cmos.
fact descriptions and family characteristics 13 2.2 capacitive loading effects in addition to temperature and power supply effects, capacitive loading effects for loads greater than 50 pf should be taken into account for propagation delays of fact devices. minimum delay numbers may be determined from table 3. propagation delays are measured to the 50% point of the output waveform. table 3 minimum propagation delay voltage(v) parameter 3.0 4.5 5.5 units t rise 31 22 19 ps/pf t fall 18 13 12.5 ps/pf note: ta = 25?c the two graphs following, figures 11 and 12, describe propagation delays on fact devices as affected by variations in power supply voltage (v cc ) and lumped load capacitance (c l ). figures 13 and 14 show the effects of lumped load capacitance on rise and fall times for fact devices. delay (ns) v cc (volts) 3.0 3.5 4.0 4.5 5.0 5.5 6.0 0 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 32 c l = 1000 pf, t plh c l = 1000 pf, t phl c l = 470 pf, t plh c l = 470 pf, t phl c l = 220 pf, t plh c l = 50 pf, t phl c l = 50 pf, t plh c l = 220 pf, t phl figure 11 propagation delay vs. v cc (hd74ac00)
fact descriptions and family characteristics 14 delay (ns) load capacitance (pf) 50 100 150 200 250 300 350 400 450 500 550 600 650 700 750 800 850 900 950 1000 0 2 4 6 8 10 12 14 16 18 20 22 t plh v cc = 4.5 v t phl v cc = 4.5 v t phl v cc = 5.0 v t phl v cc = 5.5 v t plh v cc = 5.0 v t plh v cc = 5.5 v figure 12 propagation delay vs. c l (hd74ac00) 0 1 2 3 4 5 6 7 8 9 10 t rise (ns) 10 20 30 40 50 load capacitance ( pf ) v cc = 5.0v ta = 25 c figure 13 t rise vs. capacitance
fact descriptions and family characteristics 15 0 1 2 3 4 5 6 7 8 9 10 t fall (ns) 10 20 30 40 50 load capacitance ( pf ) v cc = 5.0v ta = 25 c figure 14 t fall vs. capacitance p-channel mos +v dd -v ss n-channel mos n-substrate output input n-well n+ n+ p+ n+ p+ p+ p-well figure 15 cmos inverter cross section with latch-up circuit schematic 2.3 latch-up a major problem with cmos has been its sensitivity to latch-up, usually attributed to high parasitic gains and high input impedance (figure 15). fact logic is guaranteed not to latch-up with dynamic currents of 100 ma forced into or out of the inputs or the outputs under worst case conditions (ta = 85?c and v cc = 5.5 vdc). at room temperature the parts can typically withstand dynamic currents of over 450 ma. for most designs, latch-up will not be a problem, but the designer should be aware of its causes and how to prevent it. fact devices have been specifically designed to reduce the possibility of latch -up occurring; hitachi accomplished this by lowering the gain of the parasitic transistors, reducing n-well and p-well resistivity to increase external drive current required to cause a parasitic to turn on, and careful design and layout to minimize the substrate-injected current coupling to other circuit areas.
fact descriptions and family characteristics 16 2.4 electrostatic discharge (esd) sensitivity fact circuits show excellent resistance to esd-type damage. these logic devices are classified as category ??of mil-std-883c, test method 3015, and withstand 4000 v typically. fact logic is guaranteed to have 2000 v esd immunity on all inputs and outputs. fact parts do not require any special handling procedures. however, normal handling precautions should be observed as in the case of any semiconductor device. figure 16 shows the esd test circuit used in the sensitivity analysis for this specification. figure 17 is the pulse waveform required to perform the sensitivity test. the test procedure is as follows: five pulses, each of 2000 v, are applied to every combination of pins with a five second cool-down period between each pulse. the polarity is then reversed and the same procedure, pulse and pin combination used for an additional five discharges. continue until all pins have been tested. if none of the devices from the sample population fails the dc and ac test characteristics, the device shall be classified as category b of mil-std-883c, tm-3015. for further specifications of tm-3015, refer to the relevant standard. the voltage is increased and the testing procedure is again performed; this entire process is repeated until all pins fail. this is done to thoroughly evaluate all pins. regulated high voltage supply dc voltmeter dut waveform terminals r 1 = 800k w (min) 30 w (max) r 2 = 1500 w charge discharge high voltage relay r 3 = 1500 w calibrate test c1 = 100pf figure 16 esd test circuit
fact descriptions and family characteristics 17 100% 90% 36.8% 10% t peak t rise t decay t rise 15 ns t decay 350 ns (r 2 +r 3 )c 1 3 300 ns time voltage amptitude figure 17 esd pulse waveform
definition of specifications 18 definition of specifications 1. power dissipation-test philosophy in an erfort to reduce confusion about measuring c pd , a jedec standard test procedure (7a appendix e) has been adopted, which specifies the test setup for each type of device. this allows a device to be exercised in a consistent manner for the purpose of specification comparison. all device measurements are made with v cc = 5.0 v at 25 c, with 3-state outputs both enabled and disabled. ? gates: switch one input. bias the remaining inputs such that the output switches. ? latches: switch the enable and d inputs such that the latch toggles. ? flip-flops: switch the clock pin while changing d (or bias j and k) such that the output (s) change each clock cycle. for parts with a common clock, exercise only one flip - flop. ? decoders: switch one address pin which changes two outputs. ? multiplexers: switch one address pin with the corresponding data inputs at opposite logic levels so that the output switches. ? counters: switch the clock pin with other inputs biased such that the device counts. ? shift registers: switch the clock pin with other inputs biased such that the device counts. ? transceivers: switch one data input. for bidirectional devices enable only one direction. ? parity generator: switch one input. ? priority encoders: switch the lowest priority input. ? load capacitance: each output which is switching should be loaded with the standard 50 pf. the equivalent load capacitance, based upon the number of outputs switching and their respective frequency, is then subtracted form the measured gross c pd number to obtain the device? actual c pd value. if the device is tested at a high enough frequency, the static supply current can be ignored. thus at 1 mhz, the following formula can be used to calculate c pd : c pd = i cc / (v cc ) (1 ?10 6 ) ?equivalent load capacitance
definition of specifications 19 2. ratings and specifications table 1 absolute maximum ratings *1 parameter symbol limit unit condition supply voltage v cc ?.5 to 7.0 v dc input diode current or dc input voltage i ik ?0 ma v i = ?.5 20 ma v i = v cc +0.5 v i ?.5 to v cc +0.5 v dc output diode current or dc output voltage i ok ?0 ma v o = ?.5 50 ma v o = v cc +0.5 v o ?.5 to v cc +0.5 v dc output source or sink current i o 50 ma dc v cc or ground current per output pin i cc or i gnd 50 ma storage temperature t stg ?5 to 150 c note: 1. absolute maximum ratings are those values beyond which damage to the device may occur. obviously the databook specifications should be met, without exception to ensure that the system design is reliable over its power supply, temperature, output/input loading variables. hitachi does not recommend operation of fact circuits outside databook specifications. table 2 recommended operating conditions parameter symbol limit unit condition supply voltage (unless otherwise specified) ac devices v cc 2.0 to 6.0 v act devices 4.5 to 5.5 input voltage v i 0 to v cc v output voltage v o 0 to v cc v operating temperature t a ?0 to +85 c input rise and fall time* 1 (typical) tr, tf 150 ns/v v cc @ 3.0v (except schmitt inputs) ?c devices v in from 40 ns/v v cc @ 4.5v 30% to 70% of v cc 25 ns/v v cc @ 5.5v input rise and fall time* 1 (typical) tr, tf 10 ns/v v cc @ 4.5v (except schmitt inputs) ?ct devices v in from 8 ns/v v cc @ 5.5v 0.8 to 2.0 v, v meas from 0.8 to 2.0v note: 1. see individual data sheets for those devices which differ from the typical input rise and fall times noted here.
definition of specifications 20 table 3 dc characteristics for hd74ac family devices ta = +25 c ta = ?0 to +85 c parameter symbol v cc (v) typ guaranteed limit unit condition minimum v ih 3.0 1.5 2.1 2.1 v vout = 0.1v or high level input 4.5 2.25 3.15 3.15 v cc ?.1v voltage 5.5 2.75 3.85 3.85 maximum v il 3.0 1.5 0.9 0.9 v vout = 0.1 v or low level input 4.5 2.25 1.35 1.35 v cc ?.1v voltage 5.5 2.75 1.65 1.65 maximum v oh 3.0 2.99 2.9 2.9 v i out = ?0 m a high level 4.5 4.49 4.4 4.4 output voltage 5.5 5.49 5.4 5.4 3.0 2.58 2.48 v v in = v il or v ih 4.5 3.94 3.80 ?2 ma i oh ?4 ma 5.5 4.94 4.80 * 1 ?4 ma maximum v ol 3.0 0.002 0.1 0.1 v i out = 50 m a low level 4.5 0.001 0.1 0.1 output voltage 5.5 0.001 0.1 0.1 3.0 0.32 0.37 v v in = v il or v ih 4.5 0.32 0.37 12 ma i ol 24 ma 5.5 0.32 0.37 * 1 24 ma maximum input leakage current i in 5.5 0.1 1.0 m av i = v cc , gnd maximum 3-state current i oz 5.5 0.5 5.0 m av i (oe) = v il , v ih v i = v cc , v g nd v o = v cc , gnd minimum dynamic output current * 2 i old 5.5 86 ma v old = 1.1 v i ohd 5.5 ?5 ma v ohd = 3.85 v notes: 1. all outputs loaded; thresholds on input associated with output under test. 2. maximum test duration 2.0 ms, one output loaded at a time.
definition of specifications 21 table 4 dc characteristics for hd74act family devices ta = +25 c ta = ?0 to +85 c parameter symbol v cc (v) typ guaranteed limit unit condition minimum high v ih 4.5 1.5 2.0 2.0 v v out = 0.1v or level input 5.5 1.5 2.0 2.0 v cc ?.1v voltage maximum low v il 4.5 1.5 0.8 0.8 v v out = 0.1 v or level input 5.5 1.5 0.8 0.8 v cc ?.1v voltage maximum high v oh 4.5 4.49 4.4 4.4 v i out = ?0 m a level 5.5 5.49 5.4 5.4 output voltage 4.5 3.94 3.80 v v in = v il or v ih 5.5 4.94 4.80 i oh ?4 ma * 1 ?4 ma maximum low v ol 4.5 0.001 0.1 0.1 v i out = 50 m a level output 5.5 0.001 0.1 0.1 voltage 4.5 0.32 0.37 v v in = v il or v ih 5.5 0.32 0.37 i ol 24 ma * 1 24 ma maximum input current i in 5.5 0.1 1.0 m av i = v cc , gnd maximum 3- state current i oz 5.5 0.5 5.0 m av i = v il , v ih v o = v cc , gnd maximum i cc /input current i cct 5.5 0.6 1.5 ma v i = v cc ?2.1v minimum dynamic output current * 2 i old 5.5 86 ma v old = 1.1 v i ohd 5.5 ?5 ma v ohd = 3.85 v notes: 1. all outputs loaded; thresholds on input associated with output under test. 2. maximum test duration 2.0 ms, one output loaded at a time.
definition of specifications 22 dut test load 50pf 50 w scope 500 w 450 w open 2 ? v cc t zh t hz t zl t lz t r = 3.0 ns t f = 3.0 ns figure 1 ac loading circuit 3. ac loading and waveforms 3.1 loading circuit figure 1 shows the ac loading circuit used in characterizing and specifying propagation delays of all fact devices (hd74ac and hd74act) unless otherwise specified in the data sheet of a specific device. the use of this load, differs somewhat from previous practice provides more meaningful information and minimizes problems of instrumentation and customer correlation. in the past, +25 c propagation delays for ttl devices were specified with a load of 15 pf to ground; this required great care in building test jigs to minimize stray capacitance and implied the use of high-impedance, high-frequency scope probes. fact circuits changed to 50 pf of capacitance, allowing more leeway in stray capacitance and also loading the device during rising or falling output transitions. this more closely resembles the inloading to be expected in average applications and thus gives the designer more useful delay figures. we have incorporated this scheme into the fact product line. the net effect of the change in ac load is to increase the average observed propagation delay by about 1 ns. the 500 w resistor to ground can be a high-frequency passive probe for a sampling oscilloscope, which costs much less than the equivalent high-impedance probe. alternately, the 500 w resistor to ground can simply be a 450 w resistor feeding into a 50 w coaxial cable leading to a sampling scope input connector, with the internal 50 w termination of the scope completing the path to ground. this is the preferred scheme for correlation. (see figure 1.) with this scheme there should be a matching cable from the device input pin to the other input of the sampling scope; this also serves as a 50 w termination for the pulse generator that supplies the input signal. shown in figure 1 is a second 500 w resistor from the device output to a switch. for most measurements this switch is open; it is closed for measuring one set of the enable/disable parameters (low-to-off and off- to-low) of a 3-state output. with the switch closed, the pair of 500 w resistors and the 2 v cc supply voltage establish a quiescent high level.
definition of specifications 23 v cc 70% v cc 30% v cc 0.1v 0v v cc - 0.1v ac test input levels dc low input range dc high input range low level noise immunity high level noise immunity transition region hd74acxx devices figure 2a test input signal levels v cc 0.8v 2.0v 3.0v 0.1v 0v v cc - 0.1v ac test input levels dc low input range dc high input range low level noise immunity high level noise immunity transition region hd74actxx devices figure 2b test input signal levels (cont)
definition of specifications 24 3.2 test conditions figure 2 describes the input signal voltage levels to be used when testing fact circuits. the ac test conditions follow industry convention requiring v in to range from 0 v for a logic low to 3.0 v for a logic high for hd74act devices and 0 v to v cc for hd74ac devices. the dc parameters are normally tested with v in at guaranteed input levels, that is v ih to v il (see tables 3 and 4 for details). care must be taken to adequately decouple these high performance parts and to protect the test signals from electrical noise. in an electrically noisy environment, (e.g., a tester and handlder not specifically designed for high-speed work), dc input levels may need to be adjusted to increase the noise margin to allow for the extra noise in the tester which would not be see in a system. noise immunity testing is performed by raising v in to the nominal supply voltage of 5.0 v then dropping it to a level corresponding to v ih , and then raising it again to the 5.0 v level. noise tests can also be performed on the v il characteristics by raising v in from 0 v to v il , then returning it to 0 v. both v ih and v il noise immunity tests should not induce a switch condition on the appropriate outputs of the fact device. good high-frequency wiring practices should be used in constructing test jigs. leads on the load capacitor should be as short as possible to minimize ripples on the output waveform transitions and to minimize undershoot. generous ground metal (preferably a ground plane) should be used for the same reasons. a v cc bypass capacitor should be provided at the test socket, also with minimum lead lengths. 3.3 rise and fall times input signals should have rise and fall times of 3.0 ns and signal swing of 0 v to 3.0 v for hd74act devices or 0 v to v cc for hd74act devices. rise and fall times less than or equal to 1 ns should be used for testing f max or pulse widths. cmos devices, including, 4000 series cmos, hc, hct and fact families, tend to oscillate when the input rise and fall times become lengthy. as a direct result of its increased performance, fact devices can be more sensitive to slow input rise and fall times than other lower performance technologies. it is important to understand why this oscillation occurs. consider the outputs, where the problem is initiated. usually, cmos outputs drive capacitive loads with low dc leakage. when the output changes from a high level to a low level, or from a low level to a high level, this capacitance has to be charged or discharged. with the present high performance technologies, this charging or discharging takes place in a very short time, typically 2-3 ns. the requirement to charge or discharge the capacitive loads quickly creates a condition where the instantaneous current change through the output structure is quite high. a voltage is generated across the v cc or ground leads inside the package due to the inductance of these leads. the internal ground of the chip will change in reference to the outside world become of this induced voltage.
definition of specifications 25 consider the input. if the internal ground changes, the input voltage level appears to change to the dut. if the input rise time is slow enough, its level might still be in the device threshold region, or very close to it, when the output switches. if the internally-induced voltage is large enough, it is possible to shift the threshold region enough so that it re-crosses the input level. if the gain of the device is sufficient and the input rise or fall time is slow enough, then the device may go into oscillation. as device propagation delays become shorter, the inputs will have less time to rise or fall through the threshold region. as device gains increase, the outputs will swing more, creating more induced voltage. instantaneous current change will be greater as outputs become quicker, generating more induced voltage. package-related causes of output oscillation are not entirely to blame for problems with input rise and fall time measurements. all testers have v cc and ground leads with a finite inductance. this inductance needs to be added to the inductance in the package to determine the overall voltage which will be induced when the outputs change. as the reference for the input signals moves further away from the pin under test, the test will be more susceptible to problems caused by the inductance of the leads and stray noise. any noise on the input signal will also cause problems. with fact logic having gains as high as 100, it merely takes a 50 mv change in the input to generate a full 5 v swing on the output. 3.4 propagation delays, f max , set and hold times a 1.0 mhz square wave is recommended for most propagation delay tests (figures 3 and 4) the repetition rate must necessarily be increased for testing f max . a 50% duty cycle should always be used when testing f max . two pulse generators are usually required for testing such parameters as setup time, hold time, recovery time etc. data in data out tpxx tpxx vmi vmo vmi = 50% v cc for hd74ac devices; 1.5 v for hd74act devices vmo = 50% v cc for hd74ac/hd74act devices figure 3 waveform for inverting and non-inverting functions
definition of specifications 26 control in clock t phl t plh vmi vmi vm vmo tw t rec output vmi = 50% v cc for hd74ac devices; 1.5 v for hd74act devices vmo = 50% v cc for hd74ac/hd74act devices figure 4 propagation delay, pulse width, and t rec waveforms 3.5 enable and disable times figures 5 and 6 show that the disable times are measured at the point where the output voltage has risen or fallen by 10% from the voltage rail level (i.e., ground for t lz or v cc for t hz ) . this change enhances the repeatability of measurements, reduces test time, and gives the system designer more realistic delay times to use in calculating minimum cycle times. since the high-impedance state rising or falling waveform is rc-controlled, the first 10% of change is more linear and is less susceptible to external influences. more importantly, perhaps from the system designer? point of view, a change in voltage of 10% is adequate to ensure that a device output has turned off. measuring to a larger change in voltage merely exaggerates the apparent disable time and thus penalizes system performance since the designer must use the enable and disable times to device worst case timing signals to ensure that the output of one device is disabled befor that of another device is enabled. output control data out vmi vmo ?.3 v v cc 90% v cc t hz t zh vmi = 50% v cc for hd74ac devices; 1.5 v for hd74act devices vmo = 50% v cc for hd74ac/hd74act devices figure 5 3-state output high enable and disable times
definition of specifications 27 output control data out vmi vmo 10% v cc gnd t lz t zl vmi = 50% v cc for hd74ac devices; 1.5 v for hd74act devices vmo = 50% v cc for hd74ac/hd74act devices figure 6 3-state output low enable and disable times 3.6 electrostatic discharge precautions should be taken to prevent damage to devices by electrostatic discharge. static charge tends to accumulate on insulated surfaces such as synthetic fabrics or carpeting, plastic sheets, trays, foam, tubes or bags, and on ungrounded electrical tools or appliances. the problem is much worse in a dry atmosphere. in general, it is recommended that individuals take the precaution of touching a known ground before handling devices. more often, handling equipment that is not properly grounded, causes damage to parts. ensure that all plastic parts of the tester, that are near the device, are conductive and connected to ground. data in control input mr or clear vmi vmi vmi t su t su t h t rec vmi = 50% v cc for hd74ac devices; 1.5 v for hd74act devices vmo = 50% v cc for hd74ac/hd74act devices figure 7 setup time, hold time and recovery time
definition of specifications 28 4. symbols and terms defined for hd74ac series explanation of symbols used in electrical characteristics and recommended operating conditions table 5 dc characteristics symbol term description v ih high level input voltage high level input voltage to ensure that a logic element operates under some constraint v il low level input voltage low level input voltage to ensure that a logic element operates under some constraint v ol low level output voltage output voltage in effect when, under the input condition for bringing the output low, the rated output current is allowed to flow to the output terminal v oh high level output voltage output voltage in effect when, under the input condition for bringing the output high, the rated output current is allowed to flow to the output terminal v t + forward input threshold voltage input voltage in effect when the operation of a logic element varies as the input is allowed to go up form a voltage level lower than the reverse input threshold voltage v t - v t - reverse input threshold voltage input voltage in effect when the operation of a logic element varies as the input is allowed to drop up form a voltage level higher than the forward input threshold voltage v t + v h hysteresis voltage difference between forward input threshold voltage v t + and reverse threshold voltage v t - i in input leakage current input current that flows in when the rated maximum input voltage is applied to the input terminal i oz off-state output current (high-impedance) current that flows to the 3-state output of an element under the input condition for bringing the output to high-impedance state i cc quiescent supply current current that flows to the supply therminal (v cc ) under the rated input condition i old minimum dynamic output current output current that flows in when, under the condition for bringing the output low,the output terminal is tied to the rated out voltage v old . i ohd output current that flows out when, under the condition for bringing the output high, the output terminal is tied to the rated out voltage v ohd . i cct maximum i cc /input current that flows to the supply terminal (v cc ) under the rated input condition
definition of specifications 29 table 6 ac characteristics symbol term description f max maximum clock frequency maximum clock frequency that maintains the stable changes in output logic level in the rated sequence under the i/o condition allowing clock pulses to change the output state t tlh rise (transient) time rated time from low level to high level of a wavefrom during the defined transient period changing from low level to high level t thl fall (transient) time rated time from high level to low level of a wavefrom during the defined transient period changing from high level to low level t plh output rise propagation delay time delay time between the rated voltage levels of an i/o voltage waveform under a defined load condition, with the output changing from low level to high level t phl output fall propagation delay time delay time between the rated voltage levels of an i/o voltage waveform under a defined load condition, with the output changing from high level to low level t hz 3-state output disable time (high level) delay time between the rated voltage levels of an i/o voltage waveform under a defined load condition, with the 3-state output changing from high level to the high-impendance state t lz 3-state output disable time (low level) delay time between the rated voltage levels of an i/o voltage waveform under a defined load condition, with the 3-state output changing from low level to the high-impendance state t zh 3-state output enable time (high level) delay time between the rated voltage levels of an i/o voltage waveform under a defined load condition, with the 3-state output changing from the high-impendance state to high level t zl 3-state output enable time (low level) delay time between the rated voltage levels of an i/o voltage waveform under a defined load condition, with the 3-state output changing from the high-impendance state to low level t w pulse width duration of time between the rated levels from a leading edge to a trailing edge of a pulse waveform t h hold time time in which to hold data at the specified input terminal after a change at another related input terminal (e.g., clock input) t su setup time time in which to set up and keep data at the specified input terminal before a change at another related input terminal (e.g., clock input)
definition of specifications 30 symbol term description t rec recovery time time period between the time when data at the specified input terminal is released and the time when another related input terminal (e.g., clock input) can be changed c in input capacitance capacitance between gnd terminal and an input terminal to which 0 v is applied c pd power dissipation capacitance equivalent device power capacitance in dynamic state table 7 explanation of symbols used in function tables symbol description h high level (in steady state; written h or "h" level in sentences) l low level (in steady state; written l or "l" level in sentences) transition from l level to h level transition from h level to l level x either h or l z 3-state output off (high impedance) a? input level of steady state for each of inputs a-h q 0 q level immediately before the indicated input condition is established q 0 complement of q 0 q n q level immediately before the latest active change ( or ) occurs single h level pulse single l level pulse toggle each output is changed to the complement of the preceding state by an active input change ( or )
design considerations 31 design considerations today? system designer is faced with the problem of keeping ahead when addressing system performance and reliability. hitachi? advanced cmos helps designers achieve these goals. fact (fairchild advanced cmos technology) logic was designed to alleviate many of the drawbacks that are common to current technology logic circuits. fact logic combines the low static power consumption and the high noise margins of cmos with a high fan-out, low input loading, and a 50 w transmission line drive capability to offer a complete family of ssi and msi devices. performance features such as advanced schottky speeds at cmos power levels, advanced schottky drive, excellent noise, esd and latch-up immunity are characteristics that designers of state-of-the-art systems require. fact logic answers all of these concerns in one family of products. to fully utilize the advantages provided by fact, the system designer should have an understanding of the flexibility as well as the trade-offs of cmos design. the following section discusses common design concerns relative to the performance and requirements of fact. there are five items of interest which need to be evaluated when implementing fact devices in new designs: interfacing?interboard and technology interfaces, battery backup, and power down or live insert/ extract systems require some special thought. transmission line driving?fact has line driving capabilities superior to all cmos families and most ttl families. noise effects?as edge rates increase, the probability of crosstalk and ground bounce problems increases. the enhanced noise immunity and high threshold levels improve fact?s resistance to crosstalk problems. board layout?prudent board layout will ensure that most noise effects are minimized. power supplies and decoupling?maximize ground and v cc traces to keep v cc /ground impedance as low as possible; full ground/v cc planes are best. decouple any device driving a transmission line; otherwise add one capacitor for every package.
design considerations 32 1. interfacing fact devices have outputs which combine balanced cmos outputs with high current line driving capability. each standard output is guaranteed to source or sink 24 ma of current at worst case conditions. this allows fact circuits to drive more loads than standard advanced schottky parts; fact can directly drive als, as, ls, hc, and hct devices. ac/act nmos, cmos or ttl v cc figure 1 interfacing fact to nmos, cmos and ttl fact devices can be directly driven by both nmos and cmos families, as shown in figure 1, operating at the same rail potential without special considerations. this is possible due to the low input loading of fact product, guaranteed to be less than 1 m a per input. some older technologies, including all existing ttl families, will not be able to drive fact circuits directly; this is due to inadequate high level capability, which is guaranteed to 2.4 v. there are two simple approaches to the ttl-to-fact interface problem. a ttl-to-cmos converter can be constructed employing a resistor pull-up to v cc of approximately 4.7 k w , which is depicted in figure 2. the correct high level is seen by the cmos device while not loading down the ttl driver. ttl ac figure 2 v ih pull-up on ttl outputs unfortunately, there will be designs where including a pull-up resistor will not be acceptable. in these cases, such as a terminated ttl bus, hitachi has designed devices which offer thresholds that are ttl- compatible (figure 3). these interfaces tend to be slightly slower than their cmos-level counter-parts due to an extra buffer stage required for level conversion. ttl v cc act figure 3 ttl interfacing to hd74act
design considerations 33 ecl devices cannot directly drive fact devices. interfacing fact-to-ecl can be accomplished by using ttl-to-ecl translators and 10125 ecl-to-ttl translators in addition to following the same rules on the ttl outputs to cmos inputs (i.e., a resistor pull-up to v cc of approximately 4.7 k w ). the translation can also be accomplished by a resistive network. a three-resistor interface between fact and ecl logic is illustrated in figure 4. figures 5 and 6 show the translation from ecl-to-fact, which is somewhat more complicated. these two examples offer some possible interfaces between ecl and fact logic. ac/act +5v 560 w 510 w ?.2v 470 w figure 4 resistive fact-to-ecl translation 270 510 t1 t2 d2 d1 5.0v 510 vbb ac ecl t1,t2 - 2sc641 (k) d1,d2 - hp5082 - 2811 ?.2v figure 5 single-ended ecl-to-hd74ac circuit
design considerations 34 620 620 t1 d2 d3 d1 5.0v 510 t1,t2 - 2sc641 (k) d1,d2 - hp5082 - 2811 d3 - in414 or in 4148 ?.2v ac ecl figure 6 differential output ecl-to-hd74ac circuit it should be understood that for fact, as with other cmos technologies, input levels that are between specified input values will cause both transistors in the cmos structure to be conducting. this will cause a low resistive path from the supply rail to ground, increasing the power consumption by several orders of magnitude. it is important that cmos inputs are always driven as close as possible to the rail. figure 7 crystal oscillator circuit implemented with fact hd74ac00
design considerations 35 2. line driving with the available high-speed logic families, designers can reach new heights in system performance. yet, these faster devices require a closer look at transmission line effects. although all circuit conductors have transmission line properties, these characteristics become significant when the edge rates of the drivers are equal to or less than three times the propagation delay of the line. significant transmission line properties may be exhibited in an example where devices have edge rates of 3 ns and lines of 8 inches or greater, assuming propagation delays of 1.7 ns/ft for an unloaded printed circuit trace. of the many properties of transmission lines, two are of major interest to the system designer: zoe, the effective equivalent impedance of the line, and t pde , the effective propagation delay down the line. it should be noted that the intrinsic values of line impedance and propagation delay, z o and t pd , are geometry- dependent. once the intrinsic values are known, the effects of gate loading can be calculated. the loaded values for zoe and t pde can be calculated with: z= z 1 c / c oe o t i + t = t 1+ c / c pde pd t i where c i = intrinsic line capacitance and ct = additional capacitance due to gate loading. the formulas indicate that the loading of lines decreases the effective impedance of the line and increases the propagation delay. lines that have a propagation delay greater than one third the rise time of the signal driver should be evaluated for transmission line effects. when performing transmission line analysis on a bus, only the longest, most heavily loaded and the shortest, least loaded lines need to be analyzed. all lines in a bus should be terminated equally; if one line requires termination, all lines in the bus should be terminated. this will ensure similar signals on all of the lines.
design considerations 36 a: no termination b: series termination c: parallel termination d: ac parallel termination e: thevenin termination figure 8 termination schemes there are several termination schemes which may be used (figure 8). included are series, parallel, ac parallel and thevenin terminations. ac parallel and series terminations are the most useful for low power applications since they do not consume any dc power. parallel and thevenin terminations cause high dc power consumption. 2.1 series terminations series terminations are most useful in high-speed applications where most of the loads are at the far end of the line. loads that are between the driver and the end of the line will receive a two-step waveform. the first wave will be the incident wave. the amplitude is dependent upon the output impedance of the driver, the value of the series resistor and the impedance of the line according to the formula v w = v cc ?z oe /(z oe + r s + z s ) the amplitude will be one-half the voltage swing if r s (the series resistor) plus the output impedance (z s ) of the driver is equal to the line impedance. the second step of the waveform is the reflection from the end of the line and will have an amplitude equal to that of the first step. all devices on the line will receive a valid level only after the wave has propagated down the line and returned to the driver. therefore, all inputs will see the full voltage swing within two times the delay of the line.
design considerations 37 2.2 parallel termination parallel terminations are not generally recommended for cmos circuits due to their power consumption, which can exceed the power consumption of the logic itself. the power consumption of parallel terminations is a function of the resistor value and the duty cycle of the signal. in addition, parallel termination tends to bias the output levels of the driver towards either v cc or ground. while this feature is not desirable for driving cmos inputs, it can be useful for driving ttl inputs. 2.3 ac parallel termination ac parallel terminations work well for applications where the delays caused by series terminations are unacceptable. the effects of ac parallel terminations are similar to the effects of standard parallel terminations. the major difference is that the capacitor blocks any dc current path and helps to reduce power consumption. 2.4 thevenin termination thevenin terminations are also not generally recommended due to their power consumption. like parallel termination, a dc path to ground is created by the terminating resistors. the power consumption of a thevenin termination, though, will generally not be function of the signal duty cycle. thevenin terminations are more applicable for driving cmos inputs because they do not bias the output levels as paralleled terminations do. it should be noted that lines with thevenin terminations should not be left floating since this will cause the input levels to float between v cc or ground, increasing power consumption. fact circuits have been designed to drive 50 w transmission lines over the full temperature range. this is guaranteed by the fact family?s specified dynamic drive capability of 86 ma sink and 75 ma source current. this ensures incident wave switching on 50 w transmission lines and is consistent with the 3 ns rated edge transition time. fact devices also feature balanced output totem pole structures to allow equal source and sink current capability. this gives rise to balanced edge rates and equal rise and fall times. balanced drive capability and transition times eliminate both the need to calculate two different delay times for each signal path and the requirement to correct signal polarity for the shortest delay time. fact product inputs have been created to take full advantage of high output levels to deliver the maximum noise immunity to the system designer. v ih and v il are specified at 70% and 30% of v cc respectively. the corresponding output levels, v oh and v ol , are specified to be within 0.1 v of the rails, of which the output is sourcing or sinking 50 m a or less. these noise margins are outlined in figure 9. 70% 50% 50% 30% input thresholds figure 9 input threshold
design considerations 38 3. cmos bus loading cmos logic devices have clamp diodes from all inputs and outputs to v cc and ground. while these diodes increase system reliability by damping out undershoot and overshoot noise, they can cause problems if power is lost. figure 10 exemplifies the situation when power is removed. any input driven above the v cc pin will forward-bias the clamp diode. current can then flow into the device, and out v cc or any output that is high. depending upon the system, this current, i in , can be quite high, and may not allow the bus voltage to reach a valid high state. one possible solution to eliminate this problem is to place a series resistor in the line. v cc input output i cc i in iout figure 10 clamp diode operation 4. noise effects fact offers the best noise immunity of any competing technology available today. with input thresholds specified at 30% and 70% of v cc and outputs that drive to within 100 mv of the rails, fact devices offer noise margins approaching 30% of v cc . at 5 v v cc , fact? specified input and output levels give almost 1.5 v of noise margin for both ground and v cc -born noise. with realistic input thresholds closer to 50% of v cc , the actual margins approach 2.5 v. however, even the most advanced technology cannot alone eliminate noise problems. good circuit board layout techniques are essential to take full advantage of the superior performance of fact circuits. well-designed circuit boards also help eliminate manufacturing and testing problems. another recommended practice is to segment the board into a high-speed area, a medium-speed area and a low-speed area. the circuit areas with high current requirements (i.e., buffer circuits and high-speed logic) should be as close to the power supplies as possible; low-speed circuit areas can be furthest away. decoupling capacitors should be adjacent to all buffer chips; they should be distributed throughout the logic: one capacitor per ship. transmission lines need to be terminated to keep reflections minimal. to minimize crosstalk, long signal lines should not be close together.
design considerations 39 4.1 crosstalk the problem of crosstalk and how to deal with it is becoming more important as system performance and board densities increase. crosstalk is the capacitive coupling of signals from one line to another. the amplitude of the noise generated on the inactive line is directly related to the edge rates of the signal on the active line, the proximity of the two lines and the distance that the two lines are adjacent. crosstalk has two basic causes. forward crosstalk, figure 11, is caused by the wavefront propagating down the printed circuit trace at two different velocities. this difference in velocities is due to the difference in the dielectric constants of air (er = 1.0) and epoxy glass (er = 4.7). as the wave propagates down the trace, this difference in velocities will cause one edge to reach the end before the other. this delay is the cause of forward crosstalk; it increases with longer trace length, so consequently the magnitude of forward crosstalk will increase with distance. reverse crosstalk, figure 12, is caused by the mutual inductance and capacitance between the lines which is a transformer action. reverse crosstalk increases linearly with distance up to a critical length. this critical length is the distance that the signal can travel during its rise or fall time. although crosstalk cannot be totally eliminated, there are some design techniques that can reduce system problems resulting from crosstalk. fact? industry-leading noise margins make systems immune to crosstalk-related problems easier to design. fact? ac noise margins, shown in figures 13 and 14, exemplify the outstanding immunity to everyday noise which can effect system reliability. voltage (v) time (ns) (5.0 ns/div) key active driver forward crosstalk active receiver this figure shows traces taken on a test fixture designed to exaggerate the amplitude of crosstalk pulses. vertical scale 1.0 v/div 0.2 v/div 1.0 v/div horizontal scale 50 ns/div 5.0 ns/div 5.0 ns/div 0.0 v figure 11 forward crosstalk on pcb traces
design considerations 40 voltage (v) time (ns) (5.0 ns/div) key active driver forward crosstalk active receiver this figure shows traces taken on a test fixture designed to exaggerate the amplitude of crosstalk pulses. vertical scale 1.0 v/div 0.2 v/div 1.0 v/div horizontal scale 50 ns/div 5.0 ns/div 5.0 ns/div 0.0 v figure 12 reverse crosstalk on pcb traces gnd 30% v cc 0 0 5 10 15 20 25 pulse width (ns) 30 35 40 45 50 1 2 3 4 volts from v cc v cc = 5.0 v hazard safe figure 13 high noise margin
design considerations 41 gnd v cc 70% 0 0 5 10 15 20 25 pulse width (ns) 30 35 40 45 50 1 2 3 volts v cc = 5.0 v hazard safe figure 14 low noise margin with over 2.0 v of noise margins, the fact family offers better noise rejection than any other comparable technology. in any design, the distance that lines run adjacent to each other should be kept as short as possible. the best situation is when the lines are perpendicular to each other. for those situations where lines must run parallel, the effects of crosstalk can be minimized by line termination. terminating a line in its characteristic impedance reduces the amplitude of an initial crosstalk pulse by 50%. terminating the line will also reduce the amount of ringing (figure 15). crosstalk problems can also be reduced by moving lines further apart or by inserting ground lines or planes between them. signal on active line noise amplitude at a versus r passive line vn = coupled signal z o a r v t 100% : n o r r+z o r v 67% : r = 2z o 60% : r = 1.5z o 50% : r = z o figure 15 effects of termination on crosstalk
design considerations 42 4.2 ground bounce ground bounce occurs as a result of the intrinsic characteristics of the leadframes and bondwires of the packages used to house cmos devices. as edge rates and drive capability increase in advanced logic families, the effects of these intrinsic electrical characteristics become more pronounced. figure 16 shows a simple circuit model for a device in a leadframe driving a standard test load. the inductor l1 represents the parasitic inductance in the ground lead of the package; inductor l2 represents the parasitic inductance in the power lead of the package; inductor l3 represents the parasitic inductance in the output lead of the package; the resistor r1 represents the output impedance of the device output, and the capacitor and resistor c l and r l represent the standard test load on the output of the device. r1 l1 l2 l3 i c l r l v cc figure 16 ground bounce output model output voltage output current inductor voltage
design considerations 43 the three waveforms shown in figure 16 depict how ground bounce is generated. the first waveform shows the voltage (v) across the load as it is switched from a logic high to a logic low. the output slue rate is dependent upon the characteristics of the output transistor, the inductors l1 and l3, and c l , the load capacitance. the second waveform shows the current that is generated as the capacitor discharges [i = c l dv/dt]. the third waveform shows the voltage that is induced across the inductance in the ground lead due to the changing currents [vgb = ? ?(di/dt)]. there are many factors which affect the amplitude of the ground bounce. included are: number of outputs switching simultaneously: more outputs results in more ground bounce. type of output load: capacitive loads generate two to three times more ground bounce than typical system traces. increasing the capacitive load to approximately 60 to 70 pf increases ground bounce. beyond 70 pf, ground bounce drops off due to the filtering effect of the load. moving the load away from the output reduces the ground bounce. location of the output pin: outputs closer to the ground pin exhibit less ground bounce than those further away. voltage: lowering v cc reduces ground bounce. test fixtures: standard test fixtures generate 30 to 50% more ground bounce than a typical system since they use capacitive loads which both increase the ac load and form lcr tank circuits that oscillate. ground bounce produces several symptoms: altered device states. fact logic does not exhibit this symptom. propagation delay degraduation. fact devices are characterized not to degrade more than 250 ps per additional output switching undershoot on active outputs. the worst-case undershoot will be approximately equal to the worst-case quiet output noise. quiet output noise. fact logic?s worst-case quiet output noise has been measured to be approximately 500 to 1100 mv in actual system applications. observing either one of the following rules is sufficient to avoid running into any of the problems associated with ground bounce: first, use caution when driving asynchronous ttl-level inputs from cmos octal outputs, or second, use caution when running control lines (set, reset, load, clock, chip select) which are glitch- sensitive through the same devices that drive data or address lines. when it is not possible to avoid the above conditions, there are simple precautions available which can minimize ground bounce noise. these are: locate these outputs as close to the ground pin as possible. use the lowest v cc possible or separate the power supplies. use board design practices which reduce any additive noise sources, such as crosstalk, reflections etc.
design considerations 44 5. design rules the set of design rules listed below are recommended to ensure reliable system operation by providing the optimum power supply connection to the devices. most designers will recognize these guidelines as those they have employed with advanced bipolar logic families. use multi-layer boards with v cc and ground planes, with the device power pins soldered directly to the planes to insure the lowest power line impedances possible. use decoupling capacitors for every device, usually 0.10 m f should be adequate. these capacitors should be located as close to the ground pin as possible. avoid sockets or wirewrap boards whenever possible. do not connect capacitors from the outputs directly to ground. 5.1 decoupling requirements fact, as with other high-performance, high-drive logic families, has special decoupling and printed circuit board layout requirements. adhering to these requirements will ensure the maximum advantages are gained with fact products. local high frequency decoupling is required to supply power to the chip when it is transitioning from a low to a high value. this power is necessary to charge the load capacitance or drive a line impedance. figure 17 displays various v cc and ground layout schemes along with associated impedances. for most power distribution networks, the typical impedance is between 50 and 100 w . this impedance appears in series with the load impedance and will cause a droop in the v cc at the part. this limits the available voltage swing at the local node, unless some form of decoupling is used. this drooping of rails will cause the rise and fall times to become elongated. consider the example described in figure 18 to calculate the amount of decoupling necessary. this circuit utilizes an hd74ac240 driving a 100 w bus from a point somewhere in the middle. being in the middle of the bus, the driver will see two 100 w loads in parallel, or an effective impedance of 50 w . to switch the line from rail to rail, a drive of 78 ma is needed; more than 624 ma will be required if all eight lines switch at once. this instantaneous current requirement will generate a voltage across the impedance of the power lines, causing the actual v cc at the chip to droop. this droop limits the voltage swing available to the driver. the net effect of the voltage droop will lengthen device rise and fall times and slow system operation. a local decoupling capacitor is required to act as a low impedance supply for the driver chip during high current conditions. it will maintain the voltage with acceptable limits and keep rise and fall times to minimum. the necessary values for decoupling capacitors can be calculated with the formula given in figure 19.
design considerations 45 0.1" 0.1" 0.1" 0.04" 0.04" v cc 1/16" glass epoxy a) 50 w v cc impedance ground plane gnd v cc b) 100 w v cc impedance 1/16 board gnd v cc c) 68 w v cc impedance 1/16 board gnd v cc d) 100 w v cc impedance 1/16 board gnd v cc e) 2 w v cc impedance 032 epoxy glass figure 17 power distribution impedances 3.9 v 78 ma 4 ns ground plane worst-case octal drain = 8 78 ma = 0.624 amp. buffer output sees net 50 w load. 50 w load line on i oh -v oh characteristic shows low-to-high step of approx. 3.9 v 0 0 v i oh v out 100 w buffer data bus 1 of 8 100 w figure 18 octal buffer driving a 100 w bus q = cv i = c d v/ d t c = i d t/ d v d t = 4 10 ? i = 0.75 a c b v cc bus z cc bypass capacitors specify v cc droop = 0.1 v max. c = select c b 0.047 m f = 25 10 ? = 0.025 m f 0.1 0.624 4 10 ? v cc place one decoupling capacitor adjacent to each package driving any transmission line and distribute others evenly thoughout the logic. one capacitor per three packages. figure 19 formula for calculating decoupling capacitors in this example, if the v cc droop is to be kept below 0.1 v and the edge rate equals 4 ns, a 0.025 m f capacitor is needed.
design considerations 46 it is good practice to distribute decoupling capacitors evenly through the logic, placing one capacitor for every package. 5.2 capacitor types decoupling capacitors need to be of the high k ceramic type with low equivalent series resistance (esr), consisting primarily of series inductance and series resistance. capacitors using 5zu dielectric have suitable properties and make a good choice for decoupling capacitors; they offer minimum cost and effective performance. 6. ttl-compatible cmos designs require delta i cc consideration the fact product line is composed of two types of advanced cmos circuits: hd74ac and hd74act devices. hd74act indicates an advanced cmos device with ttl-type input thresholds for direct replacement of ls and als circuits. as this hd74act series is used to replace ttl, the delta i cct specification must be considered; this spec may be confusing and misleading to the engineer unfamiliar with cmos. it is important to understand the concept of delta i cct and how to use it within a design. first, consider where delta i cct initiates. most cmos input structures are of the totem pole type with an n-channel transistor in a series with a p-channel transistor as illustrated in figure 20. p channel n channel v cc to internal logic input pad figure 20 cmos input structure these two transistors can be modeled as variable resistors with resistances varying according to the input voltage. the resistance of an on transistor is approximately 50 w while the resistance of an off transistor is generally greater than 5 m w . when the input to this structure is at either ground or v cc , one transistor will be on and one will be off. the total series resistance of this pair will be the combination of the two individual resistances, greater than 5 m w . the leakage current will then be less than 1 m a. when the input is between ground and v cc , the resistance of the on transistor will increase while the resistance of the off transistor will decrease. the net resistance will drop due to the much larger value of the off resistance. the total series resistance can be as low as 600 w . this reduction in series resistance of the input structure will cause a corresponding increase in i cc as current flows through the input structure. figure 21 depicts typical i cc variance with input voltage for an hd74act device.
design considerations 47 1.000 0.000 0.000 5.000 0.5000v/dlc v in i cc 0.1000 ma/div figure 21 i cc versus input voltage for hd74act devices the delta i cc specification is the increase in i cc . for each input at v cc -2.1 v, the delta i cc value should be added to the quiescent supply current to arrive at the circuit? worst-case static i cc value. fortunately, there are several factors which tend to reduce the increase in i cc per input. most ttl devices will be able to drive fact inputs well beyond the ttl output specification due to fact? low input loading in a typical system. for example, fast logic outputs can drive hd74act-type inputs down to 200 mv and up to 3.5 v. additionally, the typical i cc increase per input will be less than the specified limit. as shown in the graph above, the i cc increase at v cc -2.1 v is less than 200 m a in the typical system. experiments have shown that the i cc of an hd74act240 series device typically increases only 200 m a when all of the inputs are connected to a fast device instead of ground or v cc . it is important when designing with fact, as with any ttl-compatible cmos technology, that the delta i cc specification be considered. designers should be aware of the spec?s significance and that the data book specification is a worst-case value; most systems will see values that are much less. 7. testing advanced cmos devices with i/o pins there are more and more cmos families becoming available which can replace ttl circuits. although testing these new cmos units with programs and fixtures which were developed for bipolar devices will yield acceptable results most of the time, there are some cases where this approach will cause the test engineer problems. such is the case with parts that have a bidirectional pin, exemplified by the hd74ac245/hd74act245 octal transceiver. if the proper testing methods are not followed, these types of parts may not pass those tests for i cc and input leakage currents, even when there is no fault with the devices. cmos circuits, unlike their bipolar counterparts, have static i cc specification orders of magnitude less than standard load currents. most cmos i cc specifications are usually less than 100 m a. when conducting an i cc test, greater care must be taken so that other currents will not mask the actual i cc so that other currents will not mask the actual i cc of the device. these currents are usually sourced from the inputs and outputs.
design considerations 48 since the static i cc requirements of cmos devices are so low, output load currents must be prevented from masking the current load of the device during an i cc test. even a standard 500 w load resistor will sink 10 ma at 5 v, which is more than twice the i cc level being tested. thus, most manufacturers will specify that all outputs must be unloaded during i cc tests. another area of concern is identified when considering the inputs of the device. when the input is in the transition region, i cc can be several orders of magnitude greater than the specification. when the input voltage is in the transition region, both the n-channel and the p-channel transistors in the input totem-pole structure will be slightly on, and a conduction is created from v cc to ground. this conduction path leads to the increased i cc current seen in the i cc vs. v in curve (figure 22). when the input is at either rail, the input structure no longer conducts. most i cc testing is done with all of the inputs tied to either v cc or ground. if the inputs are allowed to float, they will typically float to the middle of the transition region, and the input structure will conduct an order of magnitude more current than the actual i cc of the device under test which is being measured by the tester. when testing the i cc of a cmos hd74ac245/hd74act245, problems can arise depending upon how the test is conducted. note the structure of the hd74ac245/hd74act245?s i/o pins illustrated figure 23. 10.00 0.00 0.00 5.00 i cc (ma) v in (v) figure 22 i cc versus i in drive enable i/o pad i/o pad figure 23 hd74ac245/hd74act245 i/o structure
design considerations 49 each i/o pin is connected to both an input device and an output device. the pin can be viewed as having three states: input, output and output disabled. however, only two states actually exist. the pin is either an input or an output. when testing the i cc of the device, the pins selected as outputs by the t/r signal must either be enabled and left open or be disabled and tied to either rail. if the output device is disabled and allowed to float, the input device will also float, and an excessive amount of current will flow from v cc to ground. v cc v cc v cc input output i in i oz clamp diodes figure 24 i/o pin internal structure a simple rule to follow is to treat any output which is disable as an input. this will help insure the integrity of an i cc test. another area which might precipitate problems is the measurement of the leakages on i/o pins. the i/o pin internal structure is depicted in figure 24. the pin is internally connected to both an input device and an output device; the limit for a leakage test must be the combined i in specification of the input and the i oz specification of the output. for fact devices, i in is specified at 1 m a while i oz is specified at 5 m a. combining these gives a limit of 6 m a for i/o pins. usually, i/o pins will show leakages that are less than the i oz specification of the output alone. testing cmos circuits is no more difficult than testing their bipolar counterparts. however, there are some areas of concern that will be new to many test engineers beginning to work with cmos. becoming familiar with and understanding these areas of concern prior to creating a test philosophy will avert many problems that might otherwise arise later.
design considerations 50 8. testing disable times of 3-state outputs in a transmission line environment traditionally, the disable time of a 3-state buffer has been measured from the 50% point on the disable input, to the 10% or 90% point on the output. on a bench test site, the output waveform is generated by a load capacitor and a pull-up/pull-down resistor. this circuit gives an rc charge/discharge curve as shown in figure 25. output oe voltage (v) time (ns) 0 1 2 3 4 5 123456789101112 figure 25 typical bench 3-state waveform ate test sites generally are unable to duplicate the bench test structure. ate test loads differ because they are usually programmable and are situated away from the actual device. a commonly used test load is a wheatstone bridge. figure 26 illustraters the wheatstone bridge test structure when used on the mct 2000 test-system to duplicate the bench load. + - dut figure 26 mct wheatstone bridge test load the voltage source provides a pull-up/pull-down voltage while the current sources provide i oh and i ol . when devices with slow output slew rates are tested with the ate load, the resultant waveforms closely approximate the bench waveform, and a high degree of correlation can be achieved. however, when devices with high output slew rates are tested, different results are observed that make correlating tester results with bench results more difficult. this difference is due to the transmission line properties of the test equipment. most disable tests are preceded by establishing a current flow through the output structure. typically, these currents will be between 5 ma and 20 ma. the device is then disabled, and a comparator detects when the output has risen to the 10% or 90% level. consider the situation where the connection between the device under test (dut) and the comparator is a transmission line. visualize the device output as a switch; the effect is easier to see. there is current flowing through the line, and then the switch is opened. at the device end, the reflection coefficient changes from 0 to 1. this generates a current edge
design considerations 51 flowing back down the line equal to the current flowing in the line prior to the opening of the switch. this current wave will propagate down the line where it will encounter the high impedance tester load. this will cause the wave to be reflected back down the line toward the dut. the current wave will continue to reflect in the transmission line until it reaches the voltage applied to the tester load. at this point, the current source impedance decreases and it will dissipate the current. a typical waveshape on a modern ate is depicted in figure 27. voltage (v) output oe time (ns) 123456789101112 0 1 2 3 4 5 figure 27 typical ate 3-state waveform transmission line theory states that the voltage level of this current wave is equal to the current in the line times the impedance of the line. with typical currents as low as 5 ma and impedances of 50 to 60 w , this voltage step can be as minimal as 250 mv. if the comparator was programmed to the 10% point, it would be looking for a step of 550 mv at 5.5 v v cc . three reflections of the current pulse would be required before the comparator would detect the level. it is this added delay time caused by the transmission line environment of the ate that may cause parts to fail customers? incoming tests, even though the device meets specifications. figure 28 graphically shows this stepout. voltage (v) time (ns) 123456 b ac 7 8 9 10 11 12 0 1 2 3 4 5 figure 28 measurement stepout point a represents the typical 50% measurement point on tester driven waveforms. point b represents the point at which the delay time would be measured on a bench test fixture. point c represents where the delay time could be measured on ate fixtures. the delay time measured on the ate fixture can vary from the bench measured delay time to some greater value, depending upon the voltage level that the tester is set. if the voltage level of the tester is close to voltage levels of the plateaus, the results may become non-repeatable.

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