uc1638 uc2638 uc3638 1/98 block diagram single or dual supply operation accurate high speed oscillator differential x5 current sense amplifier bidirectional pulse-by-pulse current limiting programmable oscillator amplitude and pwm deadband dual 500ma totem pole output drivers dual 60v, 50ma open collector drivers undervoltage lockout advanced pwm motor controller features description udg-95048-4 the uc1638 family of integrated circuits are advanced pulse width modula- tors intended for a variety of pwm motor drive and amplifier applications re- quiring either uni-directional or bi-directional drive circuits. similar in architecture to the uc1637, all necessary circuitry is included to generate an analog error signal and modulate two bi-directional pulse train outputs in pro- portion to the error signal magnitude and polarity. key features of the uc1638 include a programmable high speed triangle os- cillator, a 5x differential current sensing amplifier, a high slew rate error am- plifier, high speed pwm comparators, and two 50ma open collector as well as two 500ma totem pole output stages. the individual circuit blocks are designed to provide practical operation to switching frequencies of 500khz. significant improvements in circuit speed, elimination of many external pro- gramming components, and the inclusion of a differential current sense am- plifier, allow this controller to be specified for higher performance applications, yet maintain the flexibility of the uc1637. the current sense amplifier in conjunction with the error amplifier can be configured for average current feedback. the additional open collector outputs provide a drive signal continued
2 uc1638 uc2638 uc3638 uc1638 uc2638 uc3638 absolute maximum ratings for the highside switches in a full bridge configuration. the programmable arefin pin allows for single or dual supply operation. oscillator ramp amplitude and pwm deadband are programmable by tapping a voltage di- vider off the 5v reference to the appropriate program- ming input (pvset or db). additional features include a precision externally avail- able 5v reference, undervoltage lockout, pulse-by-pulse peak current limiting, and a remote shutdown port. the uc1638 family is available in the 20 pin n, dw and j packages. consult the factory for other packaging op- tions. supply voltage vcc (referenced to vee) .............40v output drivers (aout2, bout2) currents (continuous) ........................ 0.25a currents (peak) ............................ 500ma ref output current .................... internally limited pvset, db, rt, inv, ref, csout .............0.3to10v cs+, cs- ............................. veeC1v to vcc ct, aref, arefin, comp, sd ................vee - 0.3 output voltage (aout1, bout1) ....................60v storage temperature .................... - 65c to +150c junction temperature ................... - 55c to +150c lead temperature (soldering, 10 sec.) .............. +300c currents are positive into, negative out of the specified termi- nal. consult packaging section of data book for thermal limita- tion considerations of packages. description (cont.) connection diagrams dil-20 (top view) norjpackage soic-20 (top view) dw package plcc-20 (top view) q package
3 uc1638 uc2638 uc3638 electrical characteristics unless otherwise specified; vcc = 15v, vee =C15v, ct = 680pf, rt = 3k, v pvset = 1.5v, v comp = 0v, v csout = 0v, v db = ref, v extref = 0v, v sd = vcc C 3v, t a = - 55c to 125c for the uc1638, - 25c to 85c for the uc2638, 0c to 70c for the uc3638. t a =t j . parameter test conditions min typ max units overall supply current, operating 15 23 ma uvlo threshold reference to vee 9 10 v uvlo hysteresis reference to vee 1 v voltage amplifier input offset voltage comp = 0v - 15 0 15 mv v sense bias current 0 0.5 2 m a open loop gain comp = - 5v to +5v 75 100 db cmrr v cm = - 5v to +5v 70 100 db psrr vcm = 0v, vcc - vee = 10v to 36v 70 90 db v out high inv = - 0.1v, rl = 10k 13 13.6 v v out low inv = +0.1v, rl = 10k - 13.8 C13 v slew rate rising and falling overdrive = 1v 12 v/ m s output source current comp shorted to vee 5 15 ma output sink current comp shorted to vcc 15 40 ma gain bandwidth product f in = 100khz, 10mv p-p 1 5 mhz 5v reference output voltage i ref = - 1ma, t a = 25c 4.925 5 5.075 v output voltage i ref = - 1ma 4.875 5 5.125 v load regulation i ref = - 1ma to - 10ma - 15 2 15 mv line regulation vcc - vee = 10v to 36v - 15 2 15 mv short circuit current v ref =0v 15 70 ma oscillator initial accuracy t a = 0c C 70c 86 98 110 khz voltage stability vcc - vee = 10v to 36v 2 % total variation line, temperature 76 98 120 khz pvset input bias current 0.5 3 m a pvset input voltage range (note 1) 0.5 vref v amplitude limit (note 1) vee+3 vcc - 3v aout1, bout1 output drivers output low voltage i out = 1ma, ref. to pve, pve = 0v 0.9 1.3 v i out = 50ma 1.2 1.8 v leakage current output voltage = 50v 0.1 50 m a aout2, bout2 output drivers output high voltage i out = - 20ma, ref. to pve, pve = 0v 12.2 13.5 v i out = - 100ma, ref. to pve, pve = 0v 12 13.5 v output high clamp level i out = - 20ma, ref. to pve, pve = vee 14.4 16.5 v output low voltage i out = 20ma, ref. to pve, pve = 0v 0.4 1 v i out = 100ma, ref. to pve, pve = 0v 0.6 2.2 v output rise time c out = 1nf 50 100 ns output fall time c out = 1nf 50 100 ns
4 uc1638 uc2638 uc3638 note 1: oscillator triangle amplitude = 2.5 ? pv aref. note 2: deadband = (ref - db), referenced to comp. note 3: offset = arefin - aref. aout1, bout1: aout1 and bout1 are open collector output drivers capable of sinking 50ma. these outputs can be pulled up to 60v maximum. with a few external components, these outputs can drive the opposite high side switches in a full bridge arrangement. aout2, bout2: aout2 and bout2 are totem pole output drivers capable of driving external power mos- fets directly. the peak current ratings are 500ma. an integrated zener clamp limits the drive output amplitude to approximately 14v to prevent mosfet gate oxide overstress. these outputs are configured to drive the opposite low side switches in a full bridge arrangement. aref: the voltage on aref is simply a buffered ver- sion of the voltage on arefin. in single supply applica- tions, aref should be bypassed to vee with a 0.1 m f ceramic capacitor to provide a stable reference level for the internal circuitry. arefin: the voltage on arefin is generated inter- nally by a 50% voltage divider tied between vcc and vee. as such, it provides the mid supply reference needed for the oscillator, voltage amplifier, current am- plifier and current limit comparators when operating in single supply mode. a buffer amplifier is connected be- tween arefin and aref. in bipolar supply applications arefin is usually connected to vee, which disables the buffer amplifier, and aref is connected to 0v. comp: this is the output of the high slew rate error am- plifier. the level on comp modulates the controller duty cycle via the pwm comparators and the oscillator ramp. compensation and dc gain setting resistors are con- nected between comp and inv. cs-: this is the inverting input to the x5 current sense amplifier. the common mode input range for this pin ex- tends from vee - 1v to vcc - 4v. a low value resistor in pin descriptions electrical characteristics unless otherwise specified; vcc = 15v, vee =C15v, ct = 680pf, rt = 3k, v pvset = 1.5v, v comp = 0v, v csout = 0v, v db = ref, v extref = 0v, v sd = vcc C 3v, t a = - 55c to 125c for the uc1638, - 25c to 85c for the uc2638, 0c to 70c for the uc3638. t a =t j . parameter test conditions min typ max units x5 amplifier gain v id = 100mv to 400mv 4.75 5 5.25 v/v common mode rejection v cs+ ,v cs - = aref 5v 50 65 db - 3db bandwidth 300 400 khz slew rate rising .75 1.5 v/ m s slew rate falling .75 1.5 v/ m s shutdown threshold ref. to vcc - 1.9 - 2.25 - 2.5 v input bias current v sd = sd threshold - 0.5 - 10 m a current limit threshold positive measured between cs+ and cs- 400 500 600 mv threshold negative measured between cs+ and cs- C 600 - 500 C 400 mv propagation delay to outputs overdrive = 200mv 150 250 ns deadband adjust maximum deadband v db =0v 5v zero deadband v db = ref 0 v deadband adjustment gain v db = 1v to 4v (note 2) 0.9 1 1.2 v/v input bias current v db = vref 3 15 m a aref buffer gain aref / vcc - vee 0.49 0.5 0.51 v/v offset (note 3) 30 100 mv (continued)
5 uc1638 uc2638 uc3638 pin descriptions (cont.) series with the source or emitter of the low side switch in the full bridge develops the signal that is applied to this pin. at differential inputs of 500mv typical (referenced to cs+)the controller reaches the current limit level, which truncates the output pulse. cs+: this is the non-inverting input to the x5 current sense amplifier. the common mode input range for this pin extends from vee - 1v to vcc - 4v. the characteris- tics for this pin are identical to cs-. csout: this is the output of the x5 current sense am- plifier. voltage levels greater than 2.5v referenced to aref will cause the device to enter current limit. an in- ternal 100 ohm resistor between the amplifier output and csout is provided to create a high frequency noise filter with an external capacitor to vee. when used for average current feedback, csout is summed into inv. ct: a capacitor from ct to vee will set the triangle os- cillator frequency according to the following equation: f 1 5rtct = the waveform on ct is symmetrical about the voltage on aref and is applied internally to the inputs of hte pwm comparators. use a high quality ceramic capacitor with low esl and esr for best results. a minimum ct value of 200 pf insures good accuracy and less suscep- tibility to circuit layout parasitics. the oscilator and pwm are designed to provide practical operation to 500khz. db: this high impedance input programs output pulse train deadtime. a stable dc voltage between 0v and ref will set a bi-directional deadband centered about the level on comp. the deadband level is equal to: 5v - vdb. that is, 1v on db will program 4v of deadband centered about the comp pin level. a convenient method for generating the programming level is a volt- age divider tap off of ref. inv: this is the inverting input to the voltage amplifier. the common mode input range for this pin extends from vee+2v to vcc - 1v. it can be tied to a command signal generated by a rate feedback element or to a position control signal. in average current feedback applications, this input is tied to the output of the x5 current sensing amplifier (csout). pve: this is the high current ground for the ic. the ex- ternal mosfet driver transistors are referenced to this ground. internal level shifting circuitry gives the option of tying this pin to vee, or the system ground in split sup- ply applications. pvset: a dc voltage on pvset programs the upper and lower thresholds for the oscillator by the following relationship: v pk - v vly =5?v pvset . the input voltage range on pvset is 0.5v to ref. ref: ref is the output of the precision reference. the output is capable of supplying 15ma to peripheral cir- cuitry and is internally short circuit current limited. by- pass ref to vee with a 0.1 m f ceramic capacitor for best performance. rt: a single resistor from rt to vee sets the charging and discharging currents for the triangle oscillator. the actual charge and discharge is 2x the current pro- grammed by rt and pvset. for best performance the current out of rt should be limited to 1ma. the voltage level on the rt pin is a buffered version of the pvset pin voltage. therefore, if the pvset voltage divider is tied between vcc and vee to incorporate line feedfor- ward, the triangle waveform frequency will remain con- stant. sd: a voltage on sd within 2.5v (typical) of vcc will cause the uc3638 to enter a uvlo condition which dis- ables all of the driver outputs. with an external voltage divider across vcc and vee, and a capacitor between sd and vcc, a delayed turn-on characteristic can be generated. since the 2.5v threshold is temperature sta- bilized it can also be used as a higher uvlo threshold for applications which require a starting voltage higher than the internal 9v uvlo threshold. vee: all voltages are measured with respect to this pin. all bypass capacitors and timing components except those listed under the pve section should be connected to this pin. component leads should be as short and di- rect as possible. vee is generally connected to the most negative voltage supply in the system. in single supply applications, vee is tied to the system ground. vcc: positive supply rail for the ic. bypass this pin to vee and pve with 0.1 to 1 m f low esl, esr ceramic capacitor(s). the maximum voltage for vcc is 40v ref- erenced to vee. the turn on voltage level on vcc is 9v with 1v of hysteresis.
6 uc1638 uc2638 uc3638 the uc3638 is designed to provide pulse width modula- tion control of dc brush motors in applications requiring precision torque, velocity, or position control. due to its high frequency capability, other high power applications such as switch mode audio amplifiers can also be ad- dressed. through a combination of circuit sophistication and integration, the designer can maintain a high level of flexibility, while reducing cost compared to solutions using other pwm ics. figure 1 shows a typical application circuit for the uc3638. by taking advantage of the uc3638's many in- tegrated functions, a low cost and compact average cur- rent mode motor controller can be designed. depending on the level of complexity, as many as 15 discrete com- ponents and an additional high bandwidth amplifier can be saved compared to a similar circuit using the uc3637 pwm controller. oscillator section and modulation scheme figure 2 depicts the uc3638 oscillator and pwm wave- forms for the condition where the output of the voltage amplifier (comp) is at the null point (same voltage as aref). for applications using split voltage supply rails, aref will normally be tied to system ground. this re- sults in a voltage null point of zero volts. for this condi- tion, arefin should be tied to the negative voltage sup- ply rail (vee), which disables the internal voltage buffer, allowing aref to be tied to ground. for a single supply system, arefin should be left open circuit, and aref should be decoupled to vee (system ground) with at least 0.1 m f. the resulting voltage null point for this case application information figure 1. average motor current control loop udg-95049-2 figure 2. oscillator and pwm waveforms udg-95068-1
7 uc1638 uc2638 uc3638 application information (cont.) will be half way between ground and vcc, and will auto- matically track changes in vcc. for cases where a dif- ferent null point is desired, aref can be tied to any voltage between vee + 2v and vcc - 2v. of course the user must also allow sufficient headroom for the triangle waveform. once the system null point has been chosen, the trian- gle wave amplitude and pwm deadband must be pro- grammed. the amplitude of the triangle wave is determined by trading off noise immunity and gain re- quirements. in general, the larger the triangle wave am- plitude, the greater the immunity to premature termination of pwm pulses due to switching noise. how- ever, high amplitude triangle waves require a greater voltage swing at the output of the voltage amplifier which ultimately reduces forward loop gain. programming the pwm deadband allows the user to trade off gain linearity requirements with power amplifier efficiency. if the modulator is configured as in figure 1, motor current is alternately pulsed by diagonally oppo- site drive fets when the servo loop is at null. by adjust- ing the deadband, the user can program the offset voltage at the input of the pwm comparators. this offset results in deadtime, or time when neither pwm signal is active. a minimum amount of deadtime is always recom- mended to provide cross conduction protection at the power amplifier. setting the deadtime to this minimum level will provide the maximum motor stiffness or holding torque, at the expense of power losses in the output stage. these losses result from the fact that the power amplifier is always sourcing motor current, even at null. as deadtime is increased, amplifier losses at null be- come less, at the expense of nonlinearity in the gain function. eventually, if the deadband voltage is in- creased to equal the amplitude of the triangle wave, er- ror voltages at the null point will result in no pwm pulsing, or a dead zone. after the triangle waveform am- plitude and deadband are selected, the operating fre- quency is easily set by proper selection of ct and rt. referring to figure 1, if the voltage supply rails are 15v, and the desired triangle wave oscillator amplitude is 6v p-p, pvset is set by: v pk - v vly =5?v pvset v pvset = 6 5 = 1.2v if 1v of deadband is chosen: 5 - v db =1v v db =4v in order to select the programming resistors, a source current for the reference is first selected. for a 1ma source current: r3+r4+r5= 5 isource = 5 1ma =5k r3 = 5-vdb isource = 1v 1ma =1k r4 = vdb - vpvset isource = 4v - 1.2v 1ma = 2.8k r5=5k - 1k - 2.8k = 1.2k all of the voltages described by these equations are ref- erenced to the negative supply rail. in other words, for a split supply system, vref is actually a negative voltage referenced to ground. the oscillator frequency is programmed by proper se- lection of rt and ct. if 220pf is chosen for ct, and an operating frequency of 30khz is desired, rt is chosen by: f= 1 5rtct 30khz = 1 5 220pf rt rt = 30k with rt = 30k, the charge current out of the rt pin is limited to 1.2v 30k =40 m a, which is well within the specified maximum of 1ma. to calculate the actual deadtime or minimum time be- tween pwm pulses (t db ), the ratio of the deadband voltage to the triangle wave amplitude is multiplied by half the oscillator period: t db = db vpk - vvly 1 f = 5-vdb 5 vpvset (5 rt ct) = (5 - vdb) rt ct vpvset for this example the deadtime is: t db = 1 30k 220pf 1.2 = 5.5 m sec if voltage feedforward is desired, pvset should be de- rived off of the supply rails instead of vref. this way changes in the supply voltage will linearly regulate the modulator gain, which decreases control loop suscepti- bility to line voltage variations. since the voltage on the rt pin is a buffered version of pvset, charge current tracks oscillator amplitude, and therefore the frequency
8 remains constant, preventing low frequency oscillator modulation in the presence of line voltage changes. output drivers the output driver section provides separate output driv- ers for high and low side drive of both pwm signals. for many applications, the 500ma peak output current ca- pability of the low side drivers (aout2 and bout2) is sufficient to directly connect to the appropriate low side mosfets of the h-bridge. a current limiting gate resis- tor may be used to control switching time if high levels of dv/dt or di/dt are expected at the drains of the mos- fets. if more current drive capability is required, the pwm drive signals can be buffered with bipolar transis- tors. the open collector high side drivers (aout1 and bout1) are designed to control high side p-channel mosfets. depending on voltage and speed require- ments, the driver stage can be simplified from the one shown on figure 1. if high side n-channel mosfets are desired, a boot strap or charge pump based drive circuit can be used as long as 100% duty cycle opera- tion is not required. average current control the uc3638 incorporates all of the necessary features for precise average current loop control of a dc motor. in the circuit shown in figure 1, motor current is sensed differentially across two current sense resistors.by using two current sense resistors both the current sourced from the motor voltage supply (vm) and the flyback cur- rent are sensed in the correct polarity to provide true torque control. if only one current sensed resistor is used, the flyback current will circulate through the body diodes of the lower mosfets and bypass the current sense resistor. the result will be a duty cycle dependent error term in the loop torque control function. in order to prevent high frequency spikes from contributing exces- sive error to the current control loop, the switching speed of the mosfets must be controlled so that sig- nificant transient current spikes do not couple across the drain to source capacitance of the mosfets. the x5 current amplifier multiplies the current signal by a factor of 5 and feeds the average current signal into the error amplifier. a window comparator detects if the peak current signal at the output of the current amplifier has a magnitude greater than 2.5v in either polarity and provides pulse-by-pulse peak current limiting. the loop should be designed so that peak motor current never reaches this level during normal operation. with integral compensation, the average current loop will have very high dc gain, resulting in effectively no average dc motor current error. for stability purposes, the high frequency gain of the voltage error amplifier must be designed such that magnitude of the slope of the error amplifier output (comp) must be less than or equal to the magnitude of the slope of the triangle wave- form. if rs1 = rs2 = rs, the dc gain of the current control loop can be calculated as: imotor icmd = rg2 5rg1rs if the uc3638 is set up in a simple velocity or position control loop, the feedback voltage (speed or position) is summed directly into the voltage error amplifier, and the current sense amplifier is only used for peak current limit control. the motor can also be replaced by another high power device, such as an audio speaker, and the same type of amplifier can be used. in the case of audio however, a higher switching frequency will probably be desired to prevent switching noise from infiltrating the audio frequency range. uvlo and shutdown the uc3638 contains undervoltage lockout (uvlo) cir- cuitry to prevent unwanted bridge turn-on before suffi- cient supply voltage is available. the open collector drivers (aout1 and bout1) are held off (no sink cur- rent) and the totem pole drivers (aout 2 and bout2) are pulled low until the voltage between vcc and vee reaches 9v typical. the uvlo circuitry becomes active at approximately 1v, and before this level the totem pole drivers are held low with passive pull down resistors. the shutdown pin holds the output drivers in their inac- tive state unless it is pulled 2.5v below vcc. an open collector gate or transistor can be used as an external enable signal, or a turn-on voltage higher than uvlo can be programmed with a resistive divider. in the case of figure 1, the turn on voltage v start can be calcu- lated as: v start = 2.5 (r1+ r2) r1 if a delayed start is desired, a capacitor can be placed in parallel with r1 to slow down the change in voltage at the shutdown pin, and thus provide a user programma- ble startup time. uc1638 uc2638 uc3638 application information (cont.) unitrode corporation 7 continental blvd. ? merrimack, nh 03054 tel. (603) 424-2410 ? fax (603) 424-3460
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