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1 ltc1627 monolithic synchronous step-down switching regulator figure 1b. efficiency vs output load current figure 1a. high efficiency step-down converter features descriptio n u the ltc ? 1627 is a monolithic current mode synchronous buck regulator using a fixed frequency architecture. the operating supply range is from 8.5v down to 2.65v, making it suitable for one or two lithium-ion battery-powered appli- cations. burst mode operation provides high efficiency at low load currents. 100% duty cycle provides low dropout operation, which extends operating time in battery-operated systems. the operating frequency is internally set at 350khz, allowing the use of small surface mount inductors. for switching noise sensitive applications it can be externally synchronized up to 525khz. the sync/fcb control pin guarantees regulation of secondary windings regardless of load on the main output by forcing continuous operation. burst mode operation is inhib- ited during synchronization or when the sync/fcb pin is pulled low to reduce noise and rf interference. soft-start is provided by an external capacitor. optional bootstrapping enhances the internal switch drive for single lithium-ion cell applications. the internal synchronous switch increases efficiency and eliminates the need for an external schottky diode, saving components and board space. the ltc1627 comes in an 8-lead so package. l1 15 h c out 100 f 6.3v 80.6k 249k 47pf v in 2.8v* to 8.5v v out 3.3v 1627 f01a c ss 0.1 f c in 22 f 16v 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 + + *v out connected to v in for 2.8v < v in < 3.3v output current (ma) 1 70 efficiency (%) 90 95 100 10 100 1000 1627 f01b 85 80 75 v out = 3.3v v in = 6v v in = 3.6v v in = 8.4v n high efficiency: up to 96% n constant frequency 350khz operation n 2.65v to 8.5v v in range n v out from 0.8v to v in , i out to 500ma n no schottky diode required n synchronizable up to 525khz n selectable burst mode tm operation n low dropout operation: 100% duty cycle n precision 2.5v undervoltage lockout n secondary winding regulation n current mode operation for excellent line and load transient response n low quiescent current: 200 m a n shutdown mode draws only 15 m a supply current n 1.5% reference accuracy n available in 8-lead so package applicatio n s u n cellular telephones n portable instruments n wireless modems n rf communications n distributed power systems n scanners n single and dual cell lithium typical applicatio n u , ltc and lt are registered trademarks of linear technology corporation. burst mode is a trademark of linear technology corporation.
2 ltc1627 t jmax = 125 c, q ja = 110 c/ w order part number absolute m axi m u m ratings w ww u package/order i n for m atio n w u u s8 part marking top view sync/fcb v dr v in sw i th run/ss v fb gnd s8 package 8-lead plastic so 1 2 3 4 8 7 6 5 consult factory for military grade parts. (note 1) input supply voltage ................................ C 0.3v to 10v driver supply voltage (v in C v dr ) ........... C 0.3v to 10v i th voltage .................................................. C 0.3v to 5v run/ss, v fb voltages ................................ C 0.3v to v in sync/fcb voltage ...................................... C 0.3v to v in v dr voltage (v in 5v) ............................... C 5v to 0.3v p-channel switch source current (dc) .............. 800ma n-channel switch sink current (dc) .................. 800ma peak sw sink and source current.......................... 1.5a operating ambient temperature range commercial ............................................ 0 c to 70 c industrial ........................................... C 40 c to 85 c junction temperature (note 2) ............................. 125 c storage temperature range ................. C 65 c to 150 c lead temperature (soldering, 10 sec).................. 300 c 1627 1627i t a = 25 c, v in = 5v unless otherwise specified. electrical characteristics symbol parameter conditions min typ max units i vfb feedback current (note 3) 20 60 na v fb regulated feedback voltage (note 3) l 0.788 0.80 0.812 v d v ovl d output overvoltage lockout d v ovl = v ovl C v fb 20 60 110 mv d v fb reference voltage line regulation v in = 2.8v to 8.5v (note 3) 0.002 0.01 %/v v loadreg output voltage load regulation i th sinking 2 m a (note 3) 0.5 0.8 % i th sourcing 2 m a (note 3) C 0.5 C 0.8 % i s input dc bias current (note 4) synchronized v in = 8.5v, v out = 3.3v, frequency = 525khz 450 m a burst mode operation v ith = 0v, v in = 8.5v, v sync/fcb = open 200 320 m a shutdown v run/ss = 0v, 2.65v < v in < 8.5v 15 35 m a shutdown v run/ss = 0v, v in < 2.65v 6 m a v run/ss run/ss threshold 0.4 0.7 1.0 v i run/ss soft-start current source v run/ss = 0v 1.2 2.25 3.3 m a v sync/fcb auxiliary feedback threshold v sync/fcb ramping negative 0.755 0.8 0.835 v i sync/fcb auxiliary feedback current v sync/fcb = 0v 0.5 1.5 2.5 m a f osc oscillator frequency v fb = 0.8v 315 350 385 khz v fb = 0v 35 khz v uvlo undervoltage lockout v in ramping down from 3v l 2.4 2.50 2.65 v v in ramping up from 0v l 2.65 2.80 v r pfet r ds(on) of p-channel fet (v in C v dr ) = 5v, i sw = 100ma 0.5 0.7 w r nfet r ds(on) of n-channel fet i sw = C 100ma 0.6 0.8 w i lsw sw leakage v run/ss = 0v 10 1000 na the l denotes specifications which apply over the full operating temperature range. note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: t j is calculated from the ambient temperature t a and power dissipation p d according to the following formula: t j = t a + (p d ? 110 c/w) note 3 : the ltc1627 is tested in a feedback loop that servos v fb to the balance point for the error amplifier (v ith = 0.8v). note 4: dynamic supply current is higher due to the gate charge being delivered at the switching frequency. LTC1627CS8 ltc1627is8
3 ltc1627 typical perfor a ce characteristics uw input voltage (v) 0 efficiency (%) 90 95 100 8 1627 g01 85 80 75 2 4 6 10 v out = 2.5v l = 15 m h v dr = 0v burst mode operation i load = 100ma i load = 300ma i load = 10ma efficiency vs input voltage output current (ma) 1 70 efficiency (%) 90 95 100 10 100 1000 1627 g03 85 80 75 v in = 3.6v v out = 2.5v l = 15 m h v dr = 0v burst mode operation synchronized at 525khz forced continuous output current (ma) 1 70 efficiency (%) 90 95 100 10 100 1000 1627 g02 85 80 75 v in = 3.6v v out = 2.5v l = 15 m h burst mode operation v dr = 0v v dr = v in efficiency vs load current efficiency vs load current undervoltage lockout threshold vs temperature dc supply current* vs input voltage efficiency vs load current output current (ma) 1 70 efficiency (%) 90 95 100 10 100 1000 1627 g04 85 80 75 v out = 2.5v l = 15 m h v dr = 0v burst mode operation v in = 2.8v v in = 3.6v v in = 7.2v temperature ( c) ?0 25 2.30 undervoltage lockout threshold (v) 2.35 2.45 2.50 2.55 75 100 2.75 1627 g05 2.40 0 25 50 125 2.60 2.65 2.70 v in ramping up v in ramping down input voltage (v) *does not include gate charge current 2.5 100 battery voltage (v) 150 250 300 350 6.5 550 1627 g06 200 4.5 3.5 7.5 5.5 8.5 400 450 500 t j = 25 c v out = 1.8v synchronized at 525khz burst mode operation reference voltage vs temperature forced continuous threshold voltage vs temperature supply current in shutdown vs input voltage temperature ( c) ?0 25 0.790 reference voltage (v) 0.792 0.796 0.798 0.800 75 100 0.808 1627 g08 0.794 0 25 50 125 0.802 0.804 0.806 v in = 5v temperature ( c) ?0 25 0.790 forced continuous threshold voltage (v) 0.792 0.796 0.798 0.800 75 100 0.808 1627 g09 0.794 0 25 50 125 0.802 0.804 0.806 v in = 5v input voltage (v) 2.5 4 supply current in shutdown ( m a) 6 10 12 14 6.5 22 1627 g07 8 4.5 3.5 7.5 5.5 8.5 16 18 20 v run/ss = 0v t j = 85 c t j = 25 c t j = 40 c
4 ltc1627 typical perfor a ce characteristics uw oscillator frequency vs temperature temperature ( c) ?0 25 300 oscillator frequency (khz) 310 330 340 350 75 100 390 1627 g10 320 0 25 50 125 360 370 380 v in = 5v v sync/fcb = 0v maximum output load current vs input voltage input voltage (v) 2.5 200 maximum output load current (ma) 300 500 600 700 6.5 1100 1627 g12 400 4.5 3.5 7.5 5.5 8.5 800 900 1000 v dr = ? in v dr = 0v v out = 2.5v l = 15 m h input voltage (v) 2.5 300 oscillator frequency (khz) 310 330 340 350 6.5 390 1627 g11 320 4.5 3.5 7.5 5.5 8.5 360 370 380 v sync/fcb = 0v oscillator frequency vs input voltage temperature ( c) ?0 25 0 synchronous switch leakage (na) 200 600 800 1000 75 100 1800 1627 g13 400 0 25 50 125 1200 1400 1600 v in = 8.4v v dr = 0v synchronous switch main switch switch leakage current vs temperature switch resistance vs temperature temperature ( c) ?0 25 0 switch resistance ( w ) 0.1 0.3 0.4 0.5 75 100 0.9 1627 g14 0.2 0 25 50 125 0.6 0.7 0.8 v in = 5v v dr = 0v synchronous switch main switch switch resistance vs input voltage input voltage (v) 2.5 0 switch resistance ( w ) 0.1 0.3 0.4 0.5 6.5 0.9 1627 g15 0.2 4.5 3.5 7.5 5.5 8.5 0.6 0.7 0.8 v dr = 0v synchronous switch main switch load step transient response v out 50mv/div ac coupled i th 0.5v/div i load 500ma/div 25 m s/div v in = 5v v out = 3.3v l = 15 m h c in = 22 m f c out = 100 m f i load = 0ma to 500ma burst mode operation 1627 g16 v out 20mv/div ac coupled i load 200ma/div 10 m s/div v in = 5v v out = 3.3v l = 15 m h c in = 22 m f c out = 100 m f i load = 50ma 1627 g18 v out 50mv/div ac coupled i th 0.5v/div i load 500ma/div 25 m s/div v in = 5v v out = 3.3v l = 15 m h c in = 22 m f c out = 100 m f i load = 0ma to 500ma forced continuous mode 1627 g17 load step transient response burst mode operation sw 5v/div
5 ltc1627 pi n fu n ctio n s uuu i th (pin 1): error amplifier compensation point. the current comparator threshold increases with this control voltage. nominal voltage range for this pin is 0v to 1.2v. run/ss (pin 2): combination of soft-start and run control inputs. a capacitor to ground at this pin sets the ramp time to full current output. the time is approximately 0.5s/ m f. forcing this pin below 0.4v shuts down all the circuitry. v fb (pin 3): feedback pin. receives the feedback voltage from an external resistive divider across the output. gnd (pin 4): ground pin. sw (pin 5): switch node connection to inductor. this pin connects to the drains of the internal main and synchro- nous power mosfet switches. v in (pin 6): main supply pin. must be closely decoupled to gnd, pin 4. v dr (pin 7): top driver return pin. this pin can be bootstrapped to go below ground to improve efficiency at low v in (see applications information). sync/fcb (pin 8): multifunction pin. this pin performs three functions: 1) secondary winding feedback input, 2) external clock synchronization and 3) burst mode opera- tion or forced continuous mode select. for secondary winding applications connect a resistive divider from the secondary output. to synchronize with an external clock apply a ttl/cmos compatible clock with a frequency between 385khz and 525khz. to select burst mode opera- tion, float the pin or tie it to v in . grounding pin 8 forces continuous operation (see applications information). fu n ctio n al diagra uu w v fb v in 3 2 1 6 7 5 4 q q r s 0.12v switching logic and blanking circuit 0.6v i th run/ss 0.8v 0.4v 0.86v sync/fcb y = ??only when x is a constant ?? shutdown sleep 6 1.5 a 2.25 a v in v in v in v in v dr sw gnd 1627 bd v in 0.8v burst en 8 osc freq shift 0.8v ref uvlo trip = 2.5v y x burst defeat slope comp + ovdet + + + + + + ea i comp + i rcmp run/soft start fcb anti- shoot-thru
6 ltc1627 main control loop the ltc1627 uses a constant frequency, current mode step-down architecture. both the main and synchronous switches, consisting of top p-channel and bottom n-channel power mosfets, are internal. during normal operation, the internal top power mosfet is turned on each cycle when the oscillator sets the rs latch, and turned off when the current comparator, i comp , resets the rs latch. the peak inductor current at which i comp resets the rs latch is controlled by the voltage on the i th pin, which is the output of error amplifier ea. the v fb pin, described in the pin functions section, allows ea to receive an output feedback voltage from an external resis- tive divider. when the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8v reference, which, in turn, causes the i th voltage to in- crease until the average inductor current matches the new load current. while the top mosfet is off, the bottom mosfet is turned on until either the inductor current starts to reverse as indicated by the current reversal comparator i rcmp , or the beginning of the next cycle. the main control loop is shut down by pulling the run/ss pin low. releasing run/ss allows an internal 2.25 m a current source to charge soft-start capacitor c ss . when c ss reaches 0.7v, the main control loop is enabled with the i th voltage clamped at approximately 5% of its maximum value. as c ss continues to charge, i th is gradually released, allowing normal operation to resume. comparator ovdet guards against transient overshoots > 7.5% by turning the main switch off and turning the synchronous switch on. with the synchronous switch turned on, the output is crowbarred. this may cause a large amount of current to flow from v in if the main switch is damaged, blowing the system fuse. burst mode operation the ltc1627 is capable of burst mode operation in which the internal power mosfets operate intermittently based on load demand. to enable burst mode operation, simply allow the sync/fcb pin to float or connect it to a logic high. to disable burst mode operation and enable forced continuous mode, connect the sync/fcb pin to gnd. in this mode, the efficiency is lowest at light loads, but becomes comparable to burst mode operation when the output load exceeds 100ma. the threshold voltage be- tween burst mode operation and forced continuous mode is 0.8v. this can be used to assist in secondary winding regulation as described in auxiliary winding control using sync/fcb pin in the applications information section. when the converter is in burst mode operation, the peak current of the inductor is set to approximately 200ma, even though the voltage at the i th pin indicates a lower value. the voltage at the i th pin drops when the inductors average current is greater than the load requirement. as the i th voltage drops below 0.12v, the burst comparator trips, causing the internal sleep line to go high and turn off both power mosfets. in sleep mode, both power mosfets are held off and the internal circuitry is partially turned off, reducing the quies- cent current to 200 m a. the load current is now being supplied from the output capacitor. when the output voltage drops, causing i th to rise above 0.22v, the top mosfet is again turned on and this process repeats. short-circuit protection when the output is shorted to ground, the frequency of the oscillator is reduced to about 35khz, 1/10 the nominal frequency. this frequency foldback ensures that the inductor current has more time to decay, thereby prevent- ing runaway. the oscillators frequency will progressively increase to 350khz (or the synchronized frequency) when v fb rises above 0.3v. frequency synchronization the ltc1627 can be synchronized with an external ttl/cmos compatible clock signal. the frequency range of this signal must be from 385khz to 525khz. do not attempt to synchronize the ltc1627 below 385khz as this may cause abnormal operation and an undesired fre- quency spectrum. the top mosfet turn-on follows the rising edge of the external source. when the ltc1627 is clocked by an external source, burst mode operation is disabled; the ltc1627 then operates in pwm pulse skipping mode. in this mode, when the output load is very low, current comparator i comp remains tripped for more than one cycle and forces the main switch to stay off for the same number of cycles. increasing the output operatio u (refer to functional diagram)
7 ltc1627 load slightly allows constant frequency pwm operation to resume. frequency synchronization is inhibited when the feedback voltage v fb is below 0.6v. this prevents the external clock from interfering with the frequency foldback for short- circuit protection. dropout operation when the input supply voltage decreases toward the out- put voltage, the duty cycle increases toward the maximum on-time. further reduction of the supply voltage forces the main switch to remain on for more than one cycle until it reaches 100% duty cycle. the output voltage will then be determined by the input voltage minus the voltage drop across the p-channel mosfet and the inductor. in burst mode operation or pulse skipping mode operation (externally synchronized) with the output lightly loaded, the ltc1627 transitions through continuous mode as it enters dropout. undervoltage lockout a precision undervoltage lockout shuts down the ltc1627 when v in drops below 2.5v, making it ideal for single lithium-ion battery applications. in lockout, the ltc1627 draws only several microamperes, which is low enough to prevent deep discharge and possible damage to the lithium- ion battery nearing its end of charge. a 150mv hysteresis ensures reliable operation with noisy supplies. low supply operation the ltc1627 is designed to operate down to 2.65v supply voltage. at this voltage the converter is most likely to be running at high duty cycles or in dropout where the main switch is on continuously. hence, the i 2 r loss is due mainly to the r ds(on) of the p-channel mosfet. see efficiency considerations in the applications information section. when v in is low (< 4.5v) the r ds(on) of the p-channel mosfet can be lowered by driving its gate below ground. the top p-channel mosfet driver makes use of a floating return pin, v dr , to allow biasing below gnd. a simple charge pump bootstrapped to the sw pin realizes a negative voltage at the v dr pin as shown in figure 2. using operatio u figure 3. maximum inductor peak current vs duty cycle l1 c out 100 f v out 1627 f02 v in < 4.5v d1 d2 c1 0.1 f c2 0.1 f v dr v in sw ltc1627 + 0 10 20 30 40 50 60 70 80 90 100 950 900 850 800 750 700 650 600 550 500 duty cycle (%) 1627 f03 maximum inductor peak current (ma) worst case external clock sync without external clock sync v in = 5v figure 2. using a charge pump to bias v dr the charge pump at v in 3 4.5v is not recommended to ensure that (v in C v dr ) does not exceed its absolute maximum voltage. when v in decreases to a voltage close to v out , the loop may enter dropout and attempt to turn on the p-channel mosfet continuously. when the v dr charge pump is enabled, a dropout detector counts the number of oscilla- tor cycles that the p-channel mosfet remains on, and periodically forces a brief off period to allow c1 to recharge. 100% duty cycle is allowed when v dr is grounded. slope compensation and inductor peak current slope compensation provides stability by preventing subharmonic oscillations. it works by internally adding a ramp to the inductor current signal at duty cycles in excess of 40%. as a result, the maximum inductor peak current is lower for v out /v in > 0.4 than when v out /v in < 0.4. see the inductor peak current as a function of duty cycle graph in figure 3. the worst-case peak current reduction occurs with the oscillator synchronized at its minimum frequency, i.e., to a clock just above the oscillator free-running
8 ltc1627 applicatio n s i n for m atio n wu u u frequency. the actual reduction in average current is less than for peak current. the basic ltc1627 application circuit is shown in figure 1. external component selection is driven by the load requirement and begins with the selection of l followed by c in and c out. inductor value calculation the inductor selection will depend on the operating fre- quency of the ltc1627. the internal preset frequency is 350khz, but can be externally synchronized up to 525khz. the operating frequency and inductor selection are inter- related in that higher operating frequencies allow the use of smaller inductor and capacitor values. however, oper- ating at a higher frequency generally results in lower efficiency because of internal gate charge losses. the inductor value has a direct effect on ripple current. the ripple current d i l decreases with higher inductance or frequency and increases with higher v in or v out . d i fl v v v l out out in = ()( ) - ? ? ? ? 1 1 (1) accepting larger values of d i l allows the use of low inductances, but results in higher output voltage ripple and greater core losses. a reasonable starting point for setting ripple current is d i l = 0.4(i max ). the inductor value also has an effect on burst mode operation. the transition to low current operation begins when the inductor current peaks fall to approximately 200ma. lower inductor values (higher d i l ) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. in burst mode operation, lower inductance values will cause the burst frequency to increase. inductor core selection once the value for l is known, the type of inductor must be selected. high efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy, or kool m m ? cores. actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. as inductance increases, core losses go down. unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can con- centrate on copper loss and preventing saturation. ferrite core material saturates hard, which means that induc- tance collapses abruptly when the peak design current is exceeded. this results in an abrupt increase in inductor ripple current and consequent output voltage ripple. do not allow the core to saturate! kool m m (from magnetics, inc.) is a very good, low loss core material for toroids with a soft saturation character- istic. molypermalloy is slightly more efficient at high (>200khz) switching frequencies but quite a bit more expensive. toroids are very space efficient, especially when you can use several layers of wire, while inductors wound on bobbins are generally easier to surface mount. new designs for surface mount are available from coiltronics, coilcraft and sumida. c in and c out selection in continuous mode, the source current of the top mosfet is a square wave of duty cycle v out /v in . to prevent large voltage transients, a low esr input capacitor sized for the maximum rms current must be used. the maximum rms capacitor current is given by: ci vvv v in max out in out in required i rms @ - () [] 12 / this formula has a maximum at v in = 2v out , where i rms = i out /2. this simple worst-case condition is com- monly used for design because even significant deviations do not offer much relief. note that capacitor manufacturers ripple current ratings are often based on 2000 hours of life. this makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. several capacitors may also be paralleled to meet size or height requirements in the design. always consult the manufacturer if there is any question. kool m m is a registered trademark of magnetics, inc.
9 ltc1627 applicatio n s i n for m atio n wu u u figure 4. setting the ltc1627 output voltage 0.8v v out 8.5v r2 r1 1627 f04 v fb gnd ltc1627 run/soft-start function the run/ss pin is a dual purpose pin that provides the soft-start function and a means to shut down the ltc1627. soft-start reduces surge currents from v in by gradually increasing the internal current limit. power supply sequencing can also be accomplished using this pin. an internal 2.25 m a current source charges up an external capacitor c ss . when the voltage on run/ss reaches 0.7v the ltc1627 begins operating. as the voltage on run/ss continues to ramp from 0.7v to 1.8v, the internal current limit is also ramped at a proportional linear rate. the current limit begins at 25ma (at v run/ss 0.7v) and ends at the figure 3 value (v run/ss ? 1.8v). the output current thus ramps up slowly, charging the output capacitor. if run/ss has been pulled all the way to ground, there will be a delay before the current starts increasing and is given by: t c a delay ss = 07 225 . .m pulling the run/ss pin below 0.4v puts the ltc1627 into a low quiescent current shutdown (i q < 15 m a). this pin can be driven directly from logic as shown in figure 5. diode d1 in figure 5 reduces the start delay but allows c ss to ramp up slowly providing the soft-start function. this diode can be deleted if soft-start is not needed. run/ss c ss d1 3.3v or 5v c ss run/ss 1627 f05 figure 5. run/ss pin interfacing the selection of c out is driven by the required effective series resistance (esr). typically, once the esr requirement is satisfied, the capacitance is adequate for filtering. the output ripple d v out is determined by: dd v i esr fc out l out @+ ? ? ? ? 1 4 where f = operating frequency, c out = output capacitance and d i l = ripple current in the inductor. the output ripple is highest at maximum input voltage since d i l increases with input voltage. for the ltc1627, the general rule for proper operation is: c out required esr < 0.25 w manufacturers such as nichicon, united chemicon and sanyo should be considered for high performance through- hole capacitors. the os-con semiconductor dielectric capacitor available from sanyo has the lowest esr/size ratio of any aluminum electrolytic at a somewhat higher price. once the esr requirement for c out has been met, the rms current rating generally far exceeds the i ripple(p-p) requirement. in surface mount applications multiple capacitors may have to be paralleled to meet the esr or rms current handling requirements of the application. aluminum elec- trolytic and dry tantalum capacitors are both available in surface mount configurations. in the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. an excellent choice is the avx tps series of surface mount tantalum, available in case heights ranging from 2mm to 4mm. other capacitor types include sanyo poscap, kemet t510 and t495 series, nichicon pl series and sprague 593d and 595d series. consult the manufacturer for other specific recommenda- tions. output voltage programming the output voltage is set by a resistive divider according to the following formula: vv r r out =+ ? ? ? ? 08 1 2 1 . (2) the external resistive divider is connected to the output, allowing remote voltage sensing as shown in figure 4.
10 ltc1627 auxiliary winding control using sync/fcb pin the sync/fcb pin can be used as a secondary feedback input to provide a means of regulating a flyback winding output. when this pin drops below its ground referenced 0.8v threshold, continuous mode operation is forced. in continuous mode, the main and synchronous mosfets are switched continuously regardless of the load on the main output. synchronous switching removes the normal limitation that power must be drawn from the inductor primary winding in order to extract power from auxiliary windings. with continuous synchronous operation power can be drawn from the auxiliary windings without regard to the primary output load. the secondary output voltage is set by the turns ratio of the transformer in conjunction with a pair of external resistors returned to the sync/fcb pin as shown in figure 6. the secondary regulated voltage v sec in figure 6 is given by: vnv v v r r sec out diode @+ ()( ) ->+ ? ? ? ? 1081 4 3 . where n is the turns ratio of the transformer, v out is the main output voltage sensed by v fb and v diode is the voltage drop across the schottky diode. applicatio n s i n for m atio n wu u u efficiency = 100% C (l1 + l2 + l3 + ...) where l1, l2, etc. are the individual losses as a percentage of input power. although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in ltc1627 circuits: v in quiescent current and i 2 r losses. 1. the v in quiescent current is due to two components: the dc bias current as given in the electrical character- istics and the internal main switch and synchronous switch gate charge currents. the gate charge current results from switching the gate capacitance of the internal power mosfet switches. each time the gate is switched from high to low to high again, a packet of charge dq moves from v in to ground. the resulting dq/dt is the current out of v in that is typically larger than the dc bias current. in continuous mode, i gatechg = f(q t + q b ) where q t and q b are the gate charges of the internal top and bottom switches. both the dc bias and gate charge losses are proportional to v in and thus their effects will be more pronounced at higher supply voltages. 2. i 2 r losses are calculated from the resistances of the internal switches r sw and external inductor r l . in continuous mode the average output current flowing through inductor l is chopped between the main switch and the synchronous switch. thus, the series resistance looking into sw pin from l is a function of both top and bottom mosfet r ds(on) and the duty cycle (dc) as follows: r sw = (r ds(on)top )(dc) + (r ds(on)bot )(1 C dc) the r ds(on) for both the top and bottom mosfets can be obtained from the typical performance characteris- tics curves. thus, to obtain i 2 r losses, simply add r sw to r l and multiply by the square of the average output current. other losses including c in and c out esr dissipative losses, mosfet switching losses and inductor core losses generally account for less than 2% total additional loss. figure 6. secondary output loop connection efficiency considerations the efficiency of a switching regulator is equal to the output power divided by the input power times 100%. it is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. efficiency can be expressed as: 1 f 1627 f06 r4 r3 c out v out v sec l1 1:n + + sync/fcb sw ltc1627
11 ltc1627 checking transient response the regulator loop response can be checked by looking at the load transient response. switching regulators take several cycles to respond to a step in load current. when a load step occurs, v out immediately shifts by an amount equal to ( d i load ? esr), where esr is the effective series resistance of c out . d i load also begins to charge or dis- charge c out , which generates a feedback error signal. the regulator loop then acts to return v out to its steady-state value. during this recovery time v out can be monitored for overshoot or ringing that would indicate a stability prob- lem. the internal compensation provides adequate com- pensation for most applications. but if additional compen- sation is required, the i th pin can be used for external compensation as shown in figure 7. a second, more severe transient is caused by switching in loads with large (>1 m f) supply bypass capacitors. the discharged bypass capacitors are effectively put in parallel with c out , causing a rapid drop in v out . no regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. the only solution is to limit the rise time of the switch drive so that applicatio n s i n for m atio n wu u u the load rise time is limited to approximately (25 ? c load ). thus, a 10 m f capacitor would require a 250 m s rise time, limiting the charging current to about 130ma. pc board layout checklist when laying out the printed circuit board, the following checklist should be used to ensure proper operation of the ltc1627. these items are also illustrated graphically in the layout diagram of figure 7. check the following in your layout: 1. are the signal and power grounds segregated? the ltc1627 signal ground consists of the resistive divider, the optional compensation network (r c and c c1 ), c ss and c c2 . the power ground consists of the (C) plate of c in , the (C) plate of c out and pin 4 of the ltc1627. the power ground traces should be kept short, direct and wide. the signal ground and power ground should converge to a common node in a star- ground configuration. 2. does the v fb pin connect directly to the feedback resistors? the resistive divider r1/r2 must be con- nected between the (+) plate of c out and signal ground. figure 7. ltc1627 layout diagram c c2 c c1 r c optional optional c ss c out c in d1 d2 c v c b l1 r2 r1 bold lines indicate high current paths 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 v in 1627 f07 + v out + + +
12 ltc1627 3. does the (+) plate of c in connect to v in as closely as possible? this capacitor provides the ac current to the internal power mosfets. 4. keep the switching node sw away from sensitive small- signal nodes. design example as a design example, assume the ltc1627 is used in a single lithium-ion battery-powered cellular phone applica- tion. the v in will be operating from a maximum of 4.2v down to about 2.7v. the load current requirement is a maximum of 0.5a but most of the time it will be on standby mode, requiring only 2ma. efficiency at both low and high load currents is important. output voltage is 2.5v. with this information we can calculate l using equation (1), l fi v v v l out out in = ()( ) - ? ? ? ? 1 1 d (3) substituting v out = 2.5v, v in = 4.2v, d i l = 200ma and f = 350khz in equation (3) gives: applicatio n s i n for m atio n wu u u l v khz ma v v h = ()() - ? ? ? ? = 25 350 200 1 25 42 14 5 .. . .m a 15 m h inductor works well for this application. for good efficiency choose a 1a inductor with less than 0.25 w series resistance. c in will require an rms current rating of at least 0.25a at temperature and c out will require an esr of less than 0.25 w . in most applications, the requirements for these capacitors are fairly similar. for the feedback resistors, choose r1 = 80.6k. r2 can then be calculated from equation (2) to be: r v rk out 2 08 1 1 171 =- ? ? ? ? = . ; use 169k figure 8 shows the complete circuit along with its effi- ciency curve. c ss 0.1 f c out ? 100 f 6.3v c in ?? 22 f 16v c1 0.1 f c2 0.1 f d2 d1 bat54s** 15 h* * sumida cd54-150 ** zetex bat54s ? avx tpsc107m006r0150 ?? avx tpsc226m016r0375 r2 169k 1% r1 80.6k 1% c ith 47pf 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 v in 2.8v to 4.5v v out 2.5v 0.5a 1627 f08a + + output current (ma) efficiency (%) 1 100 1000 1627 f08b 10 100 95 90 85 80 75 70 65 60 55 50 45 v out = 2.5v v in = 3.6v v in = 4.2v figure 8. single lithium-ion to 2.5v/0.5a regulator
13 ltc1627 typical applicatio n s u c ith 47pf c ss 0.1 f 15 h* 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 v in = 5v v out 3.3v 0.5a 1627 ta03 + + c out ** 100 f 6.3v c in *** 22 f 16v r2 249k 1% r1 80.6k 1% * sumida cd54-150 ** avx tpsc107m006r0150 *** avx tpsc226m016r0375 5v input to 3.3v/0.5a regulator double lithium-ion to 5v/0.5a low dropout regulator c ith 47pf c ss 0.1 f 33 h* 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 v in 8.4v v out 5v 0.5a 1627 ta04 + + c out ** 100 f 10v c in *** 22 f 16v r2 422k 1% r1 80.6k 1% * sumida cd54-330 ** avx tpsd107m010r0100 *** avx tpsc226m016r0375
14 ltc1627 typical applicatio n s u 3.3v input to 2.5v/0.5a regulator c ss 0.1 f c out ? 100 f 6.3v c in ?? 22 f 16v c1 0.1 f c2 0.1 f d2 d1 bat54s** 10 h* * sumida cd54-100 ** zetex bat54s ? avx tpsc107m006r0150 ?? avx tpsc226m016r0375 r2 169k 1% r1 80.6k 1% c ith 47pf 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 v in = 3.3v v out 2.5v 0.5a 1627 ta05 + + single lithium-ion to 1.8v/0.3a regulator c ith 47pf c ss 0.1 f 15 h* 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 v in 4.2v v out 1.8v 0.3a 1627 ta01 + + c out ** 100 f 6.3v c in *** 22 f 16v r2 100k 1% r1 80.6k 1% * sumida cd54-150 ** avx tpsc107m006r0150 *** avx tpsc226m016r0375
15 ltc1627 package descriptio n u dimensions in inches (millimeters) unless otherwise noted. 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) 0.016 ?0.050 0.406 ?1.270 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 0996 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) typ dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * ** s8 package 8-lead plastic small outline (narrow 0.150) (ltc dwg # 05-08-1610) information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
16 ltc1627 1627f lt/tp 0199 4k ? printed in usa typical applicatio n s u ? linear technology corporation 1998 double lithium-ion to 2.5v/0.5a regulator c ith 47pf c ss 0.1 f 25 h* 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 v in 8.4v v out 2.5v 0.5a 1627 ta01 + + c out ** 100 f 6.3v c in *** 22 f 16v r2 169k 1% r1 80.6k 1% * sumida cd54-250 ** avx tpsc107m006r0150 *** avx tpsc226m016r0375 ***22 m f 6.3v d1 mbr0520lt1 d2 ?? zener 1.8v * avx tpsc226m016r0375 ** avx tpsc107m006r0150 *** avx taja226m006r v sec ??? 3.3v 100ma c ith 47pf c ss 0.1 f r2 100k 1% 1627 ta02 r1 80.6k 1% v in 8.5v c in * 22 f 16v c out ** 100 f 6.3v v out 1.8v 0.3a r4 80.6k 1% r3 249k 1% 25 h ? 1:1 + + + 1 2 3 4 8 7 6 5 sync/fcb v dr v in sw i th run/ss v fb gnd ltc1627 ? coiltronics ctx25-1 ?? mmsz4678t1 ??? a 10ma min load current is recommended dual output 1.8v/300ma and 3.3v/100ma application related parts part number description comments ltc1174/ltc1174-3.3 high efficiency step-down and inverting dc/dc converters monolithic switching regulators, i out to 450ma, ltc1174-5 burst mode operation ltc1265 1.2a, high efficiency step-down dc/dc converter constant off-time, monolithic, burst mode operation lt ? 1375/lt1376 1.5a, 500khz step-down switching regulators high frequency, small inductor, high efficiency ltc1435 high efficiency, synchronous step-down converter 16-pin so and ssop ltc1436/ltc1436-pll high efficiency, low noise, synchronous step-down converters 24-pin narrow ssop ltc1438/ltc1439 dual, low noise, synchronous step-down converters multiple output capability ltc1474/ltc1475 low quiescent current step-down dc/dc converters monolithic, i out to 250ma, i q = 10 m a, 8-pin msop ltc1626 low voltage, high efficiency step-down dc/dc converter monolithic, constant off-time, i out to 600ma, low supply voltage range: 2.5v to 6v linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear-tech.com

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