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general description the max1802 provides a complete power-supply solu- tion for digital still cameras and video cameras by inte- grating two high-efficiency step-down dc-dc converters and three auxiliary step-up controllers. this complete solution is targeted for applications that use either three to four alkaline cells or two lithium-ion (li+) cells. the main step-down dc-dc controller accepts inputs from 2.5v to 11v and regulates a resistor-adjustable out- put from 2.7v to 5.5v. it uses a synchronous rectifier to regulate the output with up to 94% efficiency. an adjustable operating frequency (up to 1mhz) facilitates designs for optimum size, cost, and efficiency. the core step-down dc-dc converter accepts inputs from 2.7v to 5.5v and regulates a resistor-adjustable output from 1.25v to 5.5v. it delivers 500ma with up to 94% efficiency. the three auxiliary step-up controllers can be used to power the digital camera? ccd, lcd, and backlight. the max1802 also features expandability by supplying power, an oscillator signal, and a reference to the max1801, a low-cost slave dc-dc controller that sup- ports step-up, single-ended primary inductance con- verter (sepic), and fly-back configurations. the max1802 is available in a space-saving 32-pin tqfp package (5mm x 5mm body), and the max1801 is available in an 8-pin sot-23 package. an evaluation kit (max1802evkit) featuring both devices is available to expedite designs. ________________________applications digital still cameras digital video cameras hand-held devices internet access tablets pdas dvd players features 2.5v to 11v input voltage range main dc-dc controller 94% efficiency +2.7v to +5.5v adjustable output voltage up to 100% duty cycle independent shutdown core dc-dc converter 94% efficiency up to 500ma load efficiency output voltage adjustable down to 1.25v independent shutdown three auxiliary dc-dc controllers adjustable maximum duty cycle independent shutdown power, oscillator, and reference outputs to drive external slave controllers (max1801) up to 1mhz switching frequency 3? supply current in shutdown mode internal soft-start overload protection for all dc-dc converters compact 32-pin tqfp package max1802 digital camera step-down power supply ________________________________________________________________ maxim integrated products 1 typical operating circuit 19-1850; rev 0; 10/00 for price, delivery, and to place orders, please contact maxim distribution at 1-888-629-4642, or visit maxim? website at www.maxim-ic.com. ordering information note: refer to the separate data sheet for max1801eka in an 8- pin sot. pin configuration appears at end of data sheet. 32 tqfp pin-package temp. range -40? to +85? MAX1802EHJ part max1802 master max1801 slave input 2.5v to 11v osc power ref motor core main ccd ccfl tft
max1802 digital camera step-down power supply 2 _______________________________________________________________________________________ absolute maximum ratings electrical characteristics (circuit of figure 1, v vddm = 6v, v vddc = 3v, pgndm = pgnd = gnd, dcon1 = ref, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = 0? to +85? , unless otherwise noted. typical values are at t a = +25?.) stresses beyond those listed under ?bsolute maximum ratings?may cause permanent damage to the device. these are stress rating s only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specificatio ns is not implied. exposure to absolute maximum rating conditions for extended periods may affect device reliability. vddm, vh, onm to gnd .......................................-0.3v to +12v pgndm, pgnd to gnd ........................................-0.3v to +0.3v vh to vddm .............................................................-6v to +0.3v vl to vddm ............................................................-12v to +0.3v vl, onc, on1, fb_, dcon_ to gnd ......................-0.3v to +6v vddc, ref, osc, comp_ to gnd ..............-0.3v to (vl + 0.3v) dhm, dlm to pgndm............................-0.3v to (vddm + 0.3v) lxm to pgndm ......................................-0.6v to (vddm + 0.6v) dl1, dl2, dl3, lxc to pgnd ................-0.3v to (vddc + 0.3v) continuous power dissipation (t a = +70 c) 32-pin tqfp (derate 11.1mw/ c above +70 c)........889mw operating temperature range ...........................-40 c to +85 c junction temperature ......................................................+150 c storage temperature range. ............................-65 c to +150 c lead temperature (soldering, 10s) .................................+300 c parameter symbol conditions min typ max units general input voltage range v in 2.5 11 v supply current shutdown supply current (from vddm and vddc) v onm = 0 3 20 a v fbm = 1.5v, v vddc = 0 370 600 main dc-dc converter supply current (from vddm) v fbm = 1.5v, v vddc = 3v 35 55 a main dc-dc converter supply current (from vddc) v fbm = 1.5v, v vddc = 3v 270 450 a main plus core supply current (from vddc) v fbm = v fbc = 1.5v, v onc = 3v 410 700 a main plus auxiliary 1 supply current (from vddc) v fbm = v fb1 = 1.5v, v on1 = 3v 470 750 a main plus auxiliary 2 supply current (from vddc) v fbm = v fb2 = 1.5v, v dcon2 = 3v 470 750 a main plus auxiliary 3 supply current (from vddc) v fbm = v fb3 = 1.5v, v dcon3 = 3v 470 750 a total supply current (from vddc) v fbm = v fbc = v fb1 = v fb2 = v fb3 = 1.5v, v onc = v on1 = v dcon2 = v dcon3 = 3v 960 1700 a vl regulator vl output voltage 6v < v vddm < 11v, 0.1ma < i load < 10ma 2.83 3.00 3.12 v vl supply rejection 3.5v < v vddm < 11v, v vddc = 0 3 % vl undervoltage lockout threshold vl rising, 40mv hysteresis 2.25 2.40 2.50 v vl switchover voltage to vddc vl rising, 100mv hysteresis 2.3 2.4 2.5 v vl to vddc switch resistance 7 ?
max1802 digital camera step-down power supply _______________________________________________________________________________________ 3 electrical characteristics (continued) (circuit of figure 1, v vddm = 6v, v vddc = 3v, pgndm = pgnd = gnd, dcon1 = ref, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = 0 c to +85 c , unless otherwise noted. typical values are at t a = +25 c.) parameter symbol conditions min typ max units reference reference output voltage v ref i ref = 20 a 1.235 1.248 1.260 v ref load regulation 10 a < i ref < 200 a 5 9 mv ref line rejection 2.7v < v out < 5.5v 1 5 mv ref undervoltage lockout threshold ref rising, 20mv hysteresis 0.9 1 1.1 v oscillator osc discharge trip level osc rising 1.225 1.250 1.275 v osc input bias current v osc = 1.1v 0.2 100 na osc discharge resistance v osc = 1.5v 30 100 ? osc discharge pulse width 100 ns logic inputs (onm, onc, on1) input low level v il 0.4 v onm 1.8 input high level v ih onc, on1 1.6 v input leakage current onm: v in = 0 or 11v; onc, on1: v in = 0 or 5v 0.01 1 a main dc-dc converter main output voltage adjust range v out 2.7 5.5 v main idle mode threshold v osc = 0.625v, measured between vddm and lxm 8 20 32 mv main current-sense amplifier voltage gain a vcsm measured between vddm and lxm 8.4 9.3 10.2 v/v main n channel turn-off threshold measured between lxm and pgndm -26 -17 -8 mv main slope compensation gain a vswm 0.16 0.20 0.24 v/v main error amplifier fbm regulation voltage unity gain configuration, fbm = compm 1.233 1.248 1.263 v fbm to compm transconductance g ea unity gain configuration, fbm = compm, -5 a < i load < 5 a 70 100 160 s fbm input leakage current v fbm = 1.35v 5 100 na compm minimum output voltage v fbm = 1.35v, compm open 0.3 v compm maximum output voltage v compm ( max ) v fbm = 1.15v, compm open 2.00 2.14 2.27 v idle mode is a trademark of maxim integrated products.
max1802 digital camera step-down power supply 4 _______________________________________________________________________________________ parameter symbol conditions min typ max units main soft-start soft-start interval osc falling edge 1024 osc cycles main drivers (dhm, dlm) output low voltage i sink = 10ma 0.11 v output high voltage i source = 10ma v vddm - 0.11 v driver resistance i dhm = 10ma, i dlm = 10ma 4 11 ? drive current sourcing or sinking, v dhm or v vl = v vddm / 2 400 ma core dc-dc converter (v onc = 3v) core output voltage adjust range v out 1.25 5.5 v core idle mode threshold v osc = 0.625v 70 190 320 ma core current-sense amplifier transresistance r csc 0.7 1.0 1.3 v/a core slope compensation gain a vswc 0.16 0.20 0.24 v/v core error amplifier (v onc = 3v) fbc regulation voltage unity gain configuration, fbc = compc 1.233 1.248 1.263 v fbc to compc transconductance g ea unity gain configuration, fbc = compc, -5 a < i load < 5 a 70 100 160 s fbc input leakage current v fbc = 1.35v 5 100 na compc minimum output voltage v fbc = 1.35v, compc open 0.3 v compc maximum output voltage v compm ( max ) v fbc = 1.15v, compc open 2.00 2.14 2.27 v core soft-start (v onc = 3v) soft-start interval 1024 osc cycles core power switches (v onc = 3v) lxc leakage current v lxc = 0, 5.5v 0.01 20 a r dsn n-channel, i lxc = 0.75a 150 350 switch on-resistance r dsp p-channel, i lxc = 0.75a 180 400 m ? p-channel current limit v osc = 0.625v 0.75 a n-channel turn-off current 18 100 180 ma electrical characteristics (continued) (circuit of figure 1, v vddm = 6v, v vddc = 3v, pgndm = pgnd = gnd, dcon1 = ref, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = 0 c to +85 c , unless otherwise noted. typical values are at t a = +25 c.)
max1802 digital camera step-down power supply _______________________________________________________________________________________ 5 parameter symbol conditions min typ max units auxiliary dc-dc controllers 1, 2, 3 (v on1 = v con _ = 3v) internal clock osc clock low trip level osc falling edge 0.2 0.25 0.3 v v dcon _ = 0.625v 0.575 0.625 0.675 osc clock high trip level v dcon _ = 1.25v to v vl 1.00 1.05 1.10 v maximum duty cycle adjustment range 40 90 % maximum duty cycle v dcon _ = 0.625v 43 % default maximum duty cycle v dcon _ = 1.25v to v vl 76 % dcon_ input leakage current v dcon _ = 0v to 3v 0.01 1 a dcon_ input sleep-mode threshold v dcon _ rising, 50mv hysteresis 0.35 0.4 0.45 v auxiliary error amplifier fb_ regulation voltage unity gain configuration, fb_ = comp_ 1.233 1.248 1.263 v fb_ to comp_ transconductance g ea unity gain configuration, fb_ = comp_, -5a < iload < 5a 70 100 160 s fb_ input leakage current v fb _ = 1.35v 5 100 na auxiliary drivers (dl1, dl2, dl3) dl_ driver resistance output high or low 4 11 ? dl_ drive current sourcing or sinking, v dl _ = v vddc / 2 400 ma auxiliary soft-start soft-start interval 1024 osc cycles auxiliary short-circuit protection fault interval 1024 osc cycles electrical characteristics (continued) (circuit of figure 1, v vddm = 6v, v vddc = 3v, pgndm = pgnd = gnd, dcon1 = ref, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = 0 c to +85 c , unless otherwise noted. typical values are at t a = +25 c.)
max1802 digital camera step-down power supply 6 _______________________________________________________________________________________ electrical characteristics (circuit of figure 1, v vddm = 6v, v vddc = 3v, pgndm = pgnd = gnd, dcon1 = ref, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = -40 c to +85 c , unless otherwise noted.) (note 1) parameter symbol conditions min typ max units general input voltage range v in 2.5 11 v supply current shutdown supply current (from vddm and vddc) v onm = 0 20 a v fbm = 1.5v, v vddc = 0 600 main dc-dc converter supply current (from vddm) v fbm = 1.5v, v vddc = 3v 55 a main dc-dc converter supply current (from vddc) v fbm = 1.5v, v vddc = 3v 450 a main plus core supply current (from vddc) v fbm = v fbc = 1.5v, v onc = 3v 700 a main plus auxiliary 1 supply current (from vddc) v fbm = v fb1 = 1.5v, v on1 = v dcon1 = 3v 750 a main plus auxiliary 2 supply current (from vddc) v fbm = v fb2 = 1.5v, v dcon2 = 3v 750 a main plus auxiliary 3 supply current (from vddc) v fbm = v fb3 = 1.5v, v dcon3 = 3v 750 a total supply current (from vddc) v fbm = v fbc = v fb1 = v fb2 = v fb3 = 1.5v, v onc = v on1 = v dcon1 = v dcon2 = v dcon3 = 3v 1700 a vl regulator vl output voltage 6v < v vddm < 11v, 0.1ma < i load < 10ma 2.83 3.12 v vl supply rejection 3.5v < v vddm < 11v, v vddc = 0 3 % vl undervoltage lockout threshold v l rising, 40mv hysteresis 2.25 2.50 v vl switchover voltage to vddc v l rising, 100mv hysteresis 2.3 2.5 v vl to vddc switch resistance 7 ? reference reference output voltage v ref i ref = 20a 1.230 1.262 v ref load regulation 10a < i ref < 200a 9 mv ref line rejection 2.7v < v out < 5.5v 5 mv ref undervoltage lockout threshold ref rising, 20mv hysteresis 0.9 1.1 v oscillator osc discharge trip level osc rising 1.225 1.275 v osc input bias current v osc = 1.1v 100 na osc discharge resistance v osc = 1.5v 100 ?
max1802 digital camera step-down power supply _______________________________________________________________________________________ 7 electrical characteristics (continued) (circuit of figure 1, v vddm = 6v, v vddc = 3v, pgndm = pgnd = gnd, dcon1 = ref, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = -40 c to +85 c , unless otherwise noted.) (note 1) parameter symbol conditions min typ max units logic inputs (onm, onc, on1) input low level v il 0.4 v onm 1.8 input high level v ih onc, on1 1.6 v input leakage current onm: v in = 0 or 11v; onc, on1: v in = 0 or 5v 1 a main dc-dc converter main output voltage adjust range v out 2.7 5.5 v main idle mode threshold v osc = 0.625v, measured between vddm and lxm 2 35 mv main current-sense amplifier voltage gain a vcsm measured between vddm and lxm 8.4 10.2 v/v main zero-crossing threshold measured between lxm and pgndm -20 -8 mv main slope compensation gain a vswm 0.16 0.24 v/v main error amplifier fbm regulation voltage unity gain configuration, fbm = compm 1.230 1.265 v fbm to compm transconductance g ea u ni ty g ai n confi g ur ati on, fbm = c om p m , - 5 a < i loa d < 5 a 70 160 s fbm input leakage current v fbm = 1.35v 100 na compm minimum output voltage v fbm = 1.35v, compm open 0.3 v compm maximum output voltage v compm ( max ) v fbm = 1.15v, compm open 2.00 2.27 v main drivers (dhm, dlm) output low voltage i sink = 10ma 0.11 v output high voltage i source = 10ma v vddm - 0.11 v driver resistance i dhm = 10ma, i dlm = 10ma 11 ? core dc-dc converter (v onc = 3v) core output voltage adjust range v out 1.25 5.5 v core idle mode threshold v osc = 0.625v 40 360 ma core current-sense amplifier transresistance r csc 0.7 1.3 v/a core slope compensation gain a vswc 0.16 0.24 v/v core error amplifier (v onc = 3v) fbc regulation voltage unity gain configuration, fbc = compc 1.230 1.265 v fbc to compc transconductance g ea u ni ty g ai n confi g ur ati on, fbc = c om p c , - 5 a < i loa d < 5 a 70 160 s
max1802 digital camera step-down power supply 8 _______________________________________________________________________________________ parameter symbol conditions min typ max units fbc input leakage current v fbc = 1.35v 100 na compc minimum output voltage v fbc = 1.35v, compc open 0.3 v compc maximum output voltage v compc ( max ) v fbc = 1.15v, compc open 2.00 2.27 v core power switches (v onc = 3v) lxc leakage current v lxc = 0, 5.5v 20 a r dsn n-channel, i lxc = 0.75a 350 switch on-resistance r dsp p-channel, i lxc = 0.75a 400 m ? n-channel turn-off current 5 190 ma auxiliary dc-dc controllers 1, 2, 3 (v on1 = v dcon _= 3v) internal clock osc clock low trip level osc falling edge 0.2 0.3 v v dcon _ = 0.625v 0.575 0.675 v osc clock high trip level v dcon _ = 1.25v to v vl 1.00 1.10 maximum duty cycle adjustment range 40 90 % dcon_ input leakage current v dcon _ = 0v to 3v 1 a dcon_ input sleep-mode threshold v dcon _ rising, 50mv hysteresis 0.35 0.45 v auxiliary error amplifier fb_ regulation voltage unity gain configuration, fb_ = comp_ 1.230 1.265 v fb_ to comp_ transconductance g ea unity gain configuration, fb_ = comp_, -5a < i load < 5a 70 160 s fb_ input leakage current v fb _ = 1.35v 100 na auxiliary drivers (dl1, dl2, dl3) dl_ driver resistance output high or low 11 ? electrical characteristics (continued) (circuit of figure 1, v vddm = 6v, v vddc = 3v, pgndm = pgnd = gnd, dcon1 = ref, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = -40 c to +85 c , unless otherwise noted.) (note 1) note 1: specifications to -40 c are guaranteed by design and not production tested.
max1802 digital camera step-down power supply _______________________________________________________________________________________ 9 100 0 1 10 100 1000 efficiency vs. load current (core converter) 20 max1802 toc03 load current (ma) efficiency (%) 40 60 80 70 50 30 10 90 v in = +2.5v v in = +5v v in = +3.3v v out = +1.8v typical operating characteristics (circuit of figure 1, v vddm = 6v, v vddc = 3.3v, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = +25 c, unless otherwise noted.) 100 0 1 10 100 1000 10,000 efficiency vs. load current (main converter) 20 max1802 toc01 load current (ma) efficiency (%) 40 60 80 70 50 30 10 90 v out = 3.3v v in = +5v v in = +7.2v v in = +11v 100 0 1 10 100 1000 10,000 efficiency vs. load current (main converter) 20 max1802 toc02 load current (ma) efficiency (%) 40 60 80 70 50 30 10 90 v out = +5v v in = +7.2v v in = +11v 100 0 1 10 100 1000 efficiency vs. load current (core converter) 20 max1802 toc04 load current (ma) efficiency (%) 40 60 80 70 50 30 10 90 v in = +5v v in = +3.3v v out = +2.5v 0 20 60 40 80 100 0.4 0.6 0.7 0.5 0.8 0.9 1.0 1.1 1.2 maximum duty cycle vs. v dcon _ max1802 toc05 v dcon _ (v) maximum duty cycle (%) 0 20 60 40 80 100 0 400 200 600 800 1000 default maximum duty cycle vs. frequency max1802 toc06 frequency (khz) default maximum duty cycle (%) c osc = 470pf 1000 0 1 10 100 1000 oscillator frequency vs. r osc 200 max1802 toc07 r osc (k ? ) oscillator frequency (khz) 400 600 800 c osc = 470pf c osc = 220pf c osc = 100pf c osc = 47pf 0 2 6 4 8 10 04 2681012 max1802 toc08 input voltage (v) shutdown current ( a) shutdown current vs. input voltage
1.247 1.249 1.248 1.251 1.250 1.252 1.253 0 100 50 150 200 250 max1802 toc10 reference current ( a) reference voltage (v) reference voltage vs. reference current 0 20 10 40 30 50 60 1100 10 1000 10,000 max1802 toc11 frequency (khz) small-signal response (db) fb_ to comp_ small-signal open-loop frequency response max1802 digital camera step-down power supply 10 ______________________________________________________________________________________ typical operating characteristics (continued) (circuit of figure 1, v vddm = 6v, v vddc = 3.3v, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = +25 c, unless otherwise noted.) 1ms/div main output startup response max1802 toc12 0v 0v 0a v onm 5v/div v main 2v/div i out 200ma/div 1ms/div core output startup response max1802 toc13 0v 0v 0a v onc 5v/div v core 2v/div i out 100ma/div 1ms/div auxiliary controller startup response max1802 toc14 v on _ 5v/div v out 2v/div i out 200ma/div 0v 0v 0a 1.240 1.245 1.250 1.255 1.260 max1802 toc09 temperature ( c) reference voltage (v) -40 20 40 -20 0 60 80 reference voltage vs. temperature
max1802 digital camera step-down power supply ______________________________________________________________________________________ 11 typical operating characteristics (continued) (circuit of figure 1, v vddm = 6v, v vddc = 3.3v, v onm = 3v, v onc = v on1 = v dcon2 = v dcon3 = 0, t a = +25 c, unless otherwise noted.) 1ms/div startup sequence max1802 toc15 0v 0v 0a v onm 5v/div v main 2v/div v core 2v/div 400 s/div main output load-transient response max1802 toc16 v out ac-coupled 100mv/div i load 200ma/div 0a c out = 100 f 500 s/div core output load-transient response max1802 toc17 v out ac-coupled 200mv/div i load 100ma/div 0a v out = 2.5v 400 s/div auxiliary output load-transient response max1802 toc18 v out ac-coupled 100mv/div i load 200ma/div 0a 2.5ms/div main transient response subject to core transient max1802 toc19 v out (main) ac-coupled 20mv/div i load (core) 100ma/div 0a v out = 2.5v
max1802 digital camera step-down power supply 12 ______________________________________________________________________________________ pin description pin name function 1 fbm main dc-dc converter feedback input. connect a feedback resistive voltage-divider from the output to fbm to set the main output voltage. regulation voltage is v ref (1.25v). 2 compm compensation for main controller. output of main transconductance error amplifier. connect a series resistor and capacitor to gnd to compensate the main control loop (see compensation design ). 3 onm main converter enable input. high level turns on the main converter and vl regulator. connect onm to vddm to automatically start the converter. when the main converter is off, all other outputs are disabled. 4 vh internal bias voltage. vh provides bias to the main controller. bypass vh to vddm with a 0.1f or greater ceramic capacitor. 5 vddm battery input. vddm supplies power to the ic and also serves as a high-side current-sense input for the main dc-dc controller. connect vddm as close as possible to the source of the external p-channel switching mosfet for the main controller. 6 dhm external p-channel mosfet gate-drive output for main controller. dhm swings between vddm and pgndm with 400ma (typ) drive current. connect dhm to the gate of the external p-channel switching mosfet for the main controller. 7 lxm main dc-dc controller current-sense input. connect lxm to the drains of the external p- and n- channel switching mosfets for the main converter. lxm serves as the current-sense input for both p- and n-channel switching mosfets. connect lxm as close as possible to the drain of the external p-channel switching mosfet for the main controller. 8 dlm external n-channel mosfet gate-drive output for main controller. dlm swings between vddm and pgndm with 400ma (typ) drive current. connect dlm to the gate of the external n-channel switching mosfet for the main controller. 9 pgndm p ow er g r ound for m ai n d c - d c c ontr ol l er . p g n d m al so ser ves as a l ow - si d e cur r ent- sense i np ut for the m ai n d c - d c contr ol l er . c onnect p gn d m as cl ose as p ossi b l e to the sour ce of the exter nal n - channel sw i tchi ng m os fe t for the m ai n contr ol l er . 10 osc oscillator control. connect a timing capacitor from osc to gnd and a timing resistor from osc to vl to set the switching frequency between 100khz and 1mhz (see setting the switching frequency ). 11 dcon1 maximum duty cycle control input for auxiliary controller 1. connect dcon1 to vl to set the default maximum duty cycle. connect a resistive voltage-divider from ref to dcon1 to set the maximum duty cycle between 40% and 90%. pull dcon1 below 300mv to turn the controller off. 12 dl1 external mosfet gate drive output for auxiliary controller 1. dl1 swings between vddc and pgnd with 400ma (typ) drive current. connect dl1 to the gate of the external switching n-channel mosfet for auxiliary controller 1. 13 on1 enable input for auxiliary controller 1. connect on1 to vl to automatically start auxiliary controller 1. 14 comp1 compensation for auxiliary controller 1. output of auxiliary controller 1 transconductance error amplifier. connect a series resistor and capacitor from comp1 to gnd to compensate the auxiliary controller 1 control loop (see compensation design ). 15 fb1 feedback input for auxiliary controller 1. connect a feedback resistive voltage-divider from the output of auxiliary controller 1 to fb1 to set the output voltage. regulation voltage is v ref (1.25v). 16 fb2 feedback input for auxiliary controller 2. connect a feedback resistive voltage-divider from the output of auxiliary controller 2 to fb2 to set the output voltage. regulation voltage is v ref (1.25v).
max1802 digital camera step-down power supply ______________________________________________________________________________________ 13 pin name function 17 comp2 compensation for auxiliary controller 2. output of auxiliary controller 2 transconductance error amplifier. connect a series resistor and capacitor from comp2 to gnd to compensate the auxiliary controller 2 control loop (see compensation design ). 18 dcon2 maximum duty cycle control input for auxiliary controller 2. connect dcon2 to vl to set the default maximum duty cycle. connect a resistive voltage-divider from ref to dcon2 to set the maximum duty cycle between 40% and 90%. pull dcon2 below 300mv to turn the controller off. 19 dl2 external mosfet gate drive output for auxiliary controller 2. dl2 swings between vddc and pgnd with 400ma (typ) drive current. connect dl2 to the gate of the external switching n-channel mosfet for auxiliary controller 2. 20 dl3 external mosfet gate drive output for auxiliary controller 3. dl3 swings between vddc and pgnd with 400ma (typ) drive current. connect dl3 to the gate of the external switching n-channel mosfet for auxiliary controller 3. 21 comp3 compensation for auxiliary controller 3. output of auxiliary controller 3 transconductance error amplifier. connect a series resistor and capacitor from comp3 to gnd to compensate the auxiliary controller 3 control loop (see compensation design ). 22 fb3 feedback input for auxiliary controller 3. connect a feedback resistive voltage-divider from the output of auxiliary controller 3 to fb3 to set the output voltage. regulation voltage is v ref (1.25v). 23 dcon3 maximum duty cycle control input for auxiliary controller 3. connect dcon3 to vl to set the default maximum duty cycle. connect a resistive voltage-divider from ref to dcon3 to set the maximum duty cycle between 40% and 90%. pull dcon3 below 300mv to turn the controller off. 24 onc core converter enable input. high level turns on the core converter. connect onc to vl to automatically start the core converter. 25 pgnd power ground. sources of internal n-channel mosfet power switches. connect pgnd to gnd as close to the ic as possible. 26 lxc core power switching node. drains of the internal p- and n-channel mosfet switches for the core converter. 27 vddc core dc-dc converter power input. vddc is connected to the source of the internal p-channel mosfet power switch for the core converter. vddc is limited to 5.5v. for battery voltages greater than 5.5v, connect vddc to the main output. bypass vddc to pgnd with a 1 f or greater ceramic capacitor. 28 vl internal low-voltage bypass. the internal circuitry is powered from vl. an internal linear regulator powers vl from vddm when vddc is less than 2.4v. when vddc is greater than 2.4v, an internal switch connects vl to vddc. bypass vl to gnd with a 1.0 f or greater ceramic capacitor. 29 compc compensation for core converter. output of core transconductance error amplifier. connect a series resistor and capacitor to gnd to compensate the core control loop (see compensation design ). 30 fbc core dc-dc converter feedback input. connect a feedback resistive voltage-divider from the core output to fbc to set the output voltage. regulation voltage is v ref (1.25v). 31 ref 1.25v reference output. bypass ref to gnd with a 0.1 f or greater ceramic capacitor. 32 gnd analog ground pin description (continued)
max1802 digital camera step-down power supply 14 ______________________________________________________________________________________ detailed description the max1802 typical application circuit is shown in figure 1. it features two step-down dc-dc converters (main and core), three auxiliary step-up dc-dc con- trollers, and control capability for multiple external max1801 slave dc-dc controllers. together, these provide a complete high-efficiency power-supply solu- tion for digital still cameras. figures 2 and 3 show the max1802 functional block diagrams. master-slave configuration the max1802 supports max1801 slave controllers that obtain input power, a voltage reference, and an oscillator signal directly from the max1802 master dc-dc converter. the master-slave configuration reduces system cost by eliminating redundant circuitry and controlling the harmonic content of noise with syn- chronized converter switching. main dc-dc converter the max1802 main step-down dc-dc converter gen- erates a 2.7v to 5.5v output voltage from a 2.5v to 11v battery input voltage. when the battery voltage is lower than the main regulation voltage, the regulator goes into dropout and the p-channel switch remains on. in this condition, the output voltage is slightly lower than the input voltage. the converter drives an external p- channel mosfet power switch and an external n- channel mosfet synchronous rectifier. the converter operates in a low-noise, constant-frequency pwm cur- rent mode to regulate the voltage across the load. switching harmonics generated by fixed-frequency operation are consistent and easily filtered. the external p-channel mosfet switch turns on during the first part of each cycle, allowing current to ramp up in the inductor and store energy in a magnetic field while supplying current to the load. during the second part of each cycle, the p-channel mosfet turns off and the voltage across the inductor reverses, forcing cur- rent through the external n-channel synchronous rectifi- er to the output filter capacitor and load. as the energy stored in the inductor is depleted, the current ramps down. the synchronous rectifier turns off when the inductor current approaches zero or at the beginning of a new cycle, at which time the p-channel switch turns on again. the current-mode pwm converter uses the voltage at compm to program the inductor current and regulate the output voltage. the converter detects inductor cur- rent by sensing the voltage across the source and drain of the external p-channel mosfet. the max1802 main output switches to idle mode at light loads to improve efficiency by leaving the p-channel switch on until the voltage across the mosfet reaches the 20mv idle mode threshold. the idle mode current is 20mv divided by the mosfet on-resistance. by forcing the inductor current above the idle mode threshold, more energy is supplied to the output capacitor than is required by the load. the switch and synchronous rec- tifiers then remain off until the output capacitor dis- charges to the regulation voltage. this causes the converter to operate at a lower effective switching fre- quency at light loads, thus improving efficiency. an internal comparator turns off the n-channel synchro- nous rectifier as the inductor current drops near zero, by measuring the voltage across the mosfet. if the n- channel mosfet on-resistance is low (less than that of the p-channel switch), it may cause the mosfet to turn off prematurely, degrading efficiency. this is especially critical for high input voltage applications, such as with 2 series li+ cells. in this case, use an n-channel mos- fet with greater on-resistance than the p-channel switch, and/or place a schottky recitifier across the n- channel mosfet gate-source. the voltage at compm is typically clamped to v compm(max) = 2.14v, thereby limiting the inductor current. the peak inductor current (i lim ) and the maxi- mum average output current (i out(max) ) are deter- mined by the following equations: where a vswm is the main slope compensation gain (0.20v/v), a vcsm is the voltage gain of the main cur- rent-sense amplifier (9.3v/v), r dsp is the on-resistance of the external p-channel mosfet switch, and l is the inductor value. note that the current limit increases as the input/output voltage ratio increases. i vv va v ar ii v v v fl lim compm max ref out vswm in vcsm dsp out max lim out in out osc = ?+ ? ? ? ? ? ? =? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? () () 1 1 2
max1802 digital camera step-down power supply ______________________________________________________________________________________ 15 osc 4 5 vl vl ref ref osc gnd dcon1 dcon2 dcon3 dl3 20 dl1 d1 d2 d3 d4 q3 q4 q5 d 6 q2 dl2 fb1 12 15 fb2 19 16 dhm lxm dlm pgndm fbm vddc lxc fbc fb3 vl 28 22 23 18 11 31 10 24 13 3 2 29 14 17 21 onc on1 onm compm compc comp1 comp2 comp3 vh vddm pgnd 25 32 0.1 f 1 f 0.1 h 1 f 10 h 10 h 100 f 10 f 1 f 1 f 1 f 1 f 4.7 h 4.7 f 464k 100k 100pf 40.2k 10 f dcon d5 q1 gnd in comp dl max1802 off on 6 7 8 9 1 27 26 30 core main +12v +3.3v +1.8v lcd bias +18v -7.5v ccd bias +15v 100k 100k 100k 100k 165k 44.2k 1.34 m ? max1801 c c3 1000pf c c2 1000pf c c1 1000pf c cc 470pf c cm 4.7nf r cm 33k r cc 90k r c1 10k r c2 10k r c3 10k 2 4 3 1 7 5 6 8 c osc r osc input 2.5v to 11v +5v motor drive +7v backlight q 1 , q 2 , q 3 : fdn337n q 4 , q 5 : see mosfet selection section d 1 , d 2 , d 3 , d 4 : cmsd-4448 d 5 : mbr0502l 1.1m ? 0.1 f figure 1. typical application circuit
max1802 digital camera step-down power supply 16 ______________________________________________________________________________________ core dc-dc converter the max1802 core step-down dc-dc converter gener- ates a 1.25v to 5.5v output voltage from the main con- troller output. the core converter has the same low-noise, constant-frequency pwm current-mode architecture as the main controller. however, it uses an internal p-channel mosfet power switch and n-chan- nel mosfet synchronous rectifier to maximize efficien- cy and reduce circuit size and external component count. the core converter internally monitors the induc- tor current for current-mode regulation of the output voltage, as well as overload protection, automatic idle mode switchover, and turning off the synchronous recti- fier when the inductor current approaches zero. by switching to idle mode at light loads and turning the synchronous rectifier off at zero current, light-load effi- ciency is improved. the core converter is inactive until the main output has started. the voltage at compc is typically clamped to v compc(max) = 2.14v, thereby limiting the inductor current. the peak inductor current limit (i lim ) and the maximum average output current (i out(max) ) are determined by the following equations: where a vswc is the core slope compensation gain (0.20v/v), r csc is the transresistance of the core cur- rent-sense amplifier (1v/a), and l is the inductor value. note that the current limit increases as the input/output ratio increases. auxiliary dc-dc controllers the max1802 s three auxiliary controllers operate in a low-noise, fixed-frequency, pwm mode with output power limited by the external components. the con- i vv va v r ii v v v fl lim compc max ref out vswc in csc out max lim out in out osc = ?+ ? ? ? ? ? ? =? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? ? () () 1 1 2 osc compm fbm onm compc fbc onc soft-start v ref v ref v ref soft-start pgnd lxc vddc gnd vl pgndm dlm lxm dhm vddm vh ref reference vh main current-mode dc-dc controller vl ldo clk clk 100ns one-shot clk 2.4v core current mode dc-dc controller clock generator figure 2. simplified block diagram, including main and core
max1802 digital camera step-down power supply ______________________________________________________________________________________ 17 trollers regulate their output voltages by modulating the pulse width of the drive signal for an external n-channel mosfet switch. the auxiliary controllers are inactive until the main output has started. figure 3 shows a block diagram for a max1802 auxil- iary pwm controller. the sawtooth oscillator signal at osc governs the internal timing. at the beginning of each cycle, dl_ goes high to turn on the external mos- fet switch. the mosfet switch turns off when the internally level-shifted sawtooth rises above comp_ or when the maximum duty cycle is exceeded. the switch remains off until the beginning of the next cycle. an internal transconductance amplifier establishes an inte- grated error voltage at comp_, thereby increasing the loop gain for improved regulation accuracy. power-up sequence the max1802 is in the shutdown state with all circuitry off when the onm input is low (<1.3v). when onm goes high, an internal linear regulator generates 3v at the vl output from the vddm input to power internal circuitry. as vl rises above the 2.4v undervoltage lock- out threshold, the internal reference and oscillator begin to function and the main dc-dc converter begins soft-start operation. the main dc-dc output reaches full regulation voltage after 1024 soft-start oscillator cycles. once the main dc-dc converter com- pletes soft-start, the core dc-dc converter and the auxiliary dc-dc controllers are enabled. as the voltage at vddc rises above 2.4v, the internal linear regulator turns off and an internal 3 ? switch con- nects vl directly to vddc, which is typically connected to the output of the main dc-dc converter. the core dc-dc converter and the auxiliary dc-dc controllers have independent on-off control and soft- start. the main dc-dc converter shuts down with a low input at onm. the core dc-dc converter shuts down with a low input at onc. turn auxiliary dc-dc convert- er 1 off by driving either on1 or dcon1 to gnd. turn off auxiliary controller 2 or 3 by driving dcon2 or dcon3 to gnd. reference the max1802 has an internal 1.248v, 1% reference. connect a 0.1f bypass capacitor from ref to gnd within 0.2in (5mm) of the ref pin. ref can source up to 200a of external load current, and it is enabled whenever onm is high and vl is above the undervolt- r q dl_ s clk fault protection level shift ref comp_ fb_ dcon_ osc soft- start figure 3. auxiliary controller block diagram
max1802 digital camera step-down power supply 18 ______________________________________________________________________________________ age lockout threshold. the internal core converter, aux- iliary controllers, and max1801 slave controllers each sink up to 30a ref current during startup. if multiple max1801 controllers are turned on simultaneously, ensure that the master voltage reference can provide sufficient current, or buffer the reference with an appro- priate unity-gain amplifier. oscillator the oscillator uses a comparator, a 100ns one-shot, and an internal n-channel mosfet switch in conjunc- tion with an external timing resistor and capacitor to generate the oscillator signal at osc (figure 4). the capacitor voltage exponentially approaches vl from zero with a time constant given by the r osc c osc product when the switch is open, and the comparator output becomes high when the capacitor voltage reaches v ref (1.25v). at that time, the one-shot acti- vates the internal mosfet switch to discharge the capacitor within a 100ns interval, and the cycle repeats. note that the oscillation frequency changes as vl changes during startup. the oscillation frequency is constant while the vl voltage is constant. maximum duty cycle the max1802 s three auxiliary controllers use the saw- tooth oscillator signal generated at osc, the voltage at dcon_, and an internal comparator to limit their maxi- mum duty cycles (see setting the maximum duty cycle ). limiting the duty cycle can prevent saturation in some magnetic components. a low maximum duty cycle can also force the converter to operate in discon- tinuous current mode, simplifying design stability at the cost of a slight reduction in efficiency. soft-start all the max1802 converters feature a soft-start function that limits inrush current and prevents excessive bat- tery loading at startup by ramping the output voltage to the regulation voltage. this is achieved by increasing the internal reference inputs to the controller transcon- ductance amplifiers from 0 to the 1.25v reference volt- age over 1024 oscillator cycles when initial power is applied or when the controller is enabled. overload protection the max1802 s three auxiliary controllers have fault protection that prevents damage to transformer-cou- pled or sepic circuits due to an output overload condi- tion. when the output voltage drops out of regulation for 1024 oscillator clock periods, the auxiliary controller is disabled to prevent excessive output current. restart the controller by cycling the voltage at on_ or dcon_ to gnd and back to the on state. for a step-up appli- cation, short-circuit current is not limited, due to the dc current path through the inductor and output rectifier to the short circuit. if short-circuit protection is required in a step-up configuration, use a protection device such as a fuse to limit short-circuit current. design procedure setting the switching frequency choose a switching frequency to optimize external component size or circuit efficiency for the particular max1802 application. switching frequencies between 400khz and 500khz offer a good balance between component size and circuit efficiency. higher frequen- cies allow smaller components, and lower frequencies improve efficiency. the switching frequency is set with an external timing resistor (r osc ) and capacitor (c osc ). at the beginning of a cycle, the timing capacitor charges through the resistor until it reaches v ref . the charge time t 1 is: t 1 = -r osc (c osc +10pf) in [1 - (v ref / v vl )] once the voltage at osc reaches v ref , it discharges through an internal switch over time t 2 = 200ns. the oscillator frequency is f osc = 1 / (t 1 + t 2 ). set f osc in the range 100khz f osc 1mhz. choose c osc between 47pf and 470pf. determine r osc from the relation: r osc c osc v ref (1.25v) 100ns one-shot vl osc max1802 figure 4. oscillator
max1802 digital camera step-down power supply ______________________________________________________________________________________ 19 r osc = (200ns - 1/f osc ) / (c osc + 10pf) ? ln (1 - v ref / v vl ) see the typical operating characteristics for f osc vs. r osc using different values of c osc . due to duty cycle limitation in the main controller, keep f osc v main / (v vddm(max) ? 500ns). setting the output voltages set the max1802 output voltage of each converter by connecting a resistive voltage-divider from the output voltage to the corresponding fb_ input. the fb_ input bias current is <100na, so choose r l (the low-side fb_-to-gnd resistor) to be 100k ? . choose r h (the high-side output-to-fb_ resistor) according to the rela- tion: setting the maximum duty cycle the oscillator signal at osc and the voltage at dcon_ are used to generate the internal clock signals for the three max1802 auxiliary controllers (clk in figure 3). the internal clock s falling edge occurs when v osc exceeds v dcon _ (set by a resistive divider). the inter- nal clock s rising edge occurs when v osc falls below 0.25v (figure 5). the adjustable maximum duty cycle range is 40% to 90% (see maximum duty cycle vs. v dcon _ in the typical operating characteristics ). the maximum duty cycle defaults to 76% at 100khz if v dcon _ is at or above the voltage at v ref (1.25v) (see default maximum duty cycle vs. frequency in the typical operating characteristics ). the controller shuts down if v dcon _ is <0.3v. inductor selection main and core step-down converters max1802 main and core step-down converters offer best efficiency when the inductor current is continuous. for most designs, a reasonable inductor value (l ideal ) can be derived from the following equation, which sets continuous peak-to-peak inductor current at 1/3 the dc inductor current: where d, the duty cycle, is given by: in these equations, v dsp is the voltage drop across the p-channel mosfet switch, and v dsn is the voltage drop across the n-channel mosfet synchronous recti- fier. given l ideal , the consistent peak-to-peak inductor current is 0.33 i out . the maximum inductor current is 1.17 i out . inductance values smaller than l ideal can be used; however, the maximum inductor current will rise as l is reduced, and a larger output capacitance will be required to maintain the same output ripple. for stable operation, the minimum inductance is limited by the internal slope compensation. the minimum inductor values for main and core are given by: and where r dsp is the on-resistance of the p-channel mos- fet switch, and d max = v out / v in . auxiliary step-up controllers the three max1802 auxiliary step-up controllers offer best efficiency when the inductor current is continuous. l d v f min core max out osc () . . =? ? ? ? ? ? ? 1 05 013 l d vr f min main max out dsp osc () . . =? ? ? ? ? ? ? 1 05 0 013 d vv vv v out dsn in dsp dsn = + ?+ l vv dd if ideal in dsp out osc = ? () ? () ? ? ? ? ? ? 31 rr v hl out =? ? ? ? ? ? ? 1 248 1 . t l t h 1.25 v osc (v) v dcon_ 0.25 0 clk t h t l + t h d max = figure 5. auxiliary controller internal clock signal generation
max1802 digital camera step-down power supply 20 ______________________________________________________________________________________ use discontinuous current when the step-up ratio (v out / v in ) is greater than 1 / (1 - d max ). continuous inductor current a reasonable inductor value (l ideal ) can be derived from the following equation, which sets continuous peak-to-peak inductor current at 1/3 the dc inductor current: where d, the duty cycle, is given by: in these equations, v dsn is the voltage drop across the n-channel mosfet switch, and v d is the forward volt- age drop across the rectifier. given l ideal , the consis- tent peak-to-peak inductor current is 0.33 i out / (1 - d). the maximum inductor current is 1.17 i out / (1 - d). inductance values smaller than l ideal can be used; however, the maximum inductor current will rise as l is reduced, and a larger output capacitance will be required to maintain the same output ripple. the inductor current will become discontinuous if i out decreases by more than a factor of six from the value used to determine l ideal . discontinuous inductor current in the discontinuous mode, each max1802 auxiliary controller regulates the output voltage by adjusting the duty cycle to allow adequate power transfer to the load. to ensure regulation under worst-case load conditions (maximum i out ), choose: the peak inductor current is v in d max / (l f osc ). the inductor s saturation current rating should meet or exceed the calculated peak inductor current. input and output filter capacitors the input capacitor (c in ) reduces the current peaks drawn from the battery or input power source. the impedance of the input capacitor at the switching fre- quency should be less than that of the input source so that high-frequency switching currents do not pass through the input source. the output capacitor is required to keep the output volt- age ripple small and to ensure regulation control-loop stability. the output capacitor must have low imped- ance at the switching frequency. tantalum and ceramic capacitors are good choices. tantalum capacitors typi- cally have high capacitance and medium-to-low equiv- alent series resistance (esr) so that esr dominates the impedance at the switching frequency. in turn, the out- put ripple is approximately: v ripple i l ( p-p) esr where i l (p-p) is the peak-to-peak inductor current. ceramic capacitors typically have lower esr than tan- talum capacitors, but with relatively small capacitance that dominates the impedance at the switching fre- quency. in turn, the output ripple is approximately: v ripple i l ( p-p) z c where i l (p-p) is the peak-to-peak inductor current, and z c 1 / (2 f osc c out ). see the compensation design section for a discussion of the influence of output capacitance and esr on reg- ulation control-loop stability. the capacitor voltage rating must exceed the maximum applied capacitor voltage. for most tantalum capaci- tors, manufacturers suggest derating the capacitor by applying no more than 70% of the rated voltage to the capacitor. ceramic capacitors are typically used up to the voltage rating of the capacitor. consult the manu- facturer s specifications for proper capacitor derating. mosfet selection the max1802 main converter and auxiliary controllers drive external logic-level p- and/or n-channel mosfets as the circuit switching elements. the key selection parameters are: on-resistance (r ds(on) ) maximum drain-to-source voltage (v ds(max) ) total gate charge (q g ) reverse transfer capacitance (c rss ) because the main converter s external mosfets are used for current sense, they directly determine the out- put current capability and efficiency of the main con- verter. it is important to select the appropriate external mosfets for the main converter. the p-channel on- resistance (r dsp ) at minimum input voltage (v vddm ) must be low enough so that the converter can produce the desired output current as determined by the i out(max) equation in the main dc-dc converter sec- tion. the n-channel on-resistance (r dsn ) determines l vd if out max out osc = 2 d v vv in out d ? + 1 l vvdd if ideal in max dsn out osc = ? () ? () () 31
max1802 digital camera step-down power supply ______________________________________________________________________________________ 21 the n-channel turn-off current (equal to 17mv/r dsn ). choose r dsn value between r dsp and 3r dsp to keep the n-channel turn-off current low for optimal efficiency. if a lower r dsn is used, connect a schottky diode from pgndm to lxm for better efficiency (see diode selection ). for the main converter, the external gate drive swings between the voltage at vddm and gnd. for the auxil- iary controllers, the external gate drive swings between the voltage at vddc and gnd. use a mosfet whose on-resistance is specified at or below the minimum gate drive voltage swing, and make sure that the maxi- mum voltage swing does not exceed the maximum gate-source voltage specification of the mosfet. the gate charge, q g , includes all capacitance associated with gate charging and helps to predict the transition time required to drive the mosfet between on and off states. the power dissipated in the mosfet is due to r ds(on) and transition losses. the r ds(on) loss is: p 1 d i l 2 r ds(on) where d is the duty cycle, i l is the average inductor current, and r ds(on) is the on-resistance of the mos- fet. the transition loss is approximately: where v swing is v out for the auxiliary controllers or v in(max) for the main and core converters, i l is the average inductor current, f osc is the converter switch- ing frequency, and t t is the transition time. the transi- tion time is approximately q g / i g , where q g is the total gate charge, and i g is the gate drive current (0.4a typ). the total power dissipation in the mosfet is p mosfet = p 1 + p 2 . diode selection the main and core converters use synchronous recti- fiers and thus do not require a diode. however, if the external n-channel synchronous rectifier has low on- resistance (less than the p-channel on-resistance), the high n-channel turn-off current results in lower efficien- cy. in that case, connect a schottky diode, rated for maximum output current, from pgndm to lxm to improve efficiency. the auxiliary controllers require external rectifiers. for low-output-voltage applications, use a schottky diode to rectify the output voltage because of the diode s low forward voltage and fast recovery time. schottky diodes exhibit significant leakage current at high reverse volt- ages and high temperatures. thus, for high-voltage, high-temperature applications, use ultra-fast junction rectifiers. compensation design each dc-dc converter has an internal transconduc- tance error amplifier whose output is used to compen- sate the control loop. typically, a series resistor and capacitor are inserted from comp_ to gnd to form a pole-zero pair. the external inductor, the output capac- itor, the compensation resistor and capacitor, and for the main converter, the external p-channel mosfet, govern control-loop stability. the inductor and output capacitor are usually chosen in consideration of perfor- mance, size, and cost, but the compensation resistor and capacitor are chosen to optimize control-loop sta- bility. the component values in the circuit of figure 1 yield stable operation over a broad range of input/out- put voltages and converter switching frequencies. follow the procedures below for optimal compensation. in the following descriptions, bode plots are used to graphically describe the loop response of the convert- ers over frequency. the bode plot shows loop gain and phase vs. frequency. a single pole results in a -20db per decade slope and a -90 phase shift, and a single zero results in a +20db per decade slope and a +90 phase shift. the stability of the system can be deter- mined by the phase margin (how far from 0 the loop phase is when the response drops to 0db) and gain margin (how far below 0db the gain is when the phase reaches 0 ). the system is stable for phase margins >30 , and a phase margin of 45 is preferred. the gain margin should be at least 10db. main converter the main converter uses current mode to regulate the output voltage by forcing the required current through the inductor. since the p-channel mosfet operates with constant drain-source on-resistance (r dsp ), the voltage across the mosfet is proportional to the inductor current. the converter current-sense amplifier measures the on mosfet drain-source voltage to determine the inductor current for regulation. the gain through the current-sense amplifier (measured across the mosfet) is a vcsm = 9.3v/v. the voltage-divider attenuates the loop gain by a vdv = v ref / v out , and the gain dc voltage of the error amplifier is a vea = 2000v/v. the controller forces the peak inductor cur- rent (i l ) such that: i l r dsp a vcsm = v out a vdv a vea or i l = v out a vdv a vea / (a vcsm r dsp ) p vift swing l osc t 2 3
max1802 digital camera step-down power supply 22 ______________________________________________________________________________________ and the output voltage is i out r load , which is equal to i l r load . thus, the total dc loop gain is: a vdc = r load a vdv a vea / (a vcsm r dsp ) or a vdc = 215 v ref r load / (v out r ds(on) ) because of the current-mode control, there is a single pole in the loop response due to the output capacitor. this pole is at the frequency (in hz): p o = 1 / (2 r load c out ) note that as the load resistance increases, the pole moves to a lower frequency. however, the dc loop gain increases by the same amount since they are both dependent on r load . thus, the crossover frequency (frequency at which the loop gain drops to 0db), which is the product of the pole and the gain, remains at the same frequency. the compensation network creates a pole and zero at the frequencies (in hz): p c = g ea / (4000 c c ) = 1 / (4x10 7 c c ) and z c = 1 / (2 r c c c ) and the esr of the output filter capacitor causes a zero in the loop response at the frequency (in hz): z o = 1 / (2 c out esr) the dc gain and the poles and zeros are shown in the bode plot of figure 6. to achieve a stable circuit with the bode plot of figure 6, use the following procedure: 1) determine the desired crossover frequency, either 1/3 of the zero due to the output capacitor esr: or 1/5 of the switching frequency: whichever is lower. 2) determine the pole frequency due to the output capacitor and the load resistor: or 3) determine the compensation resistor required to set the desired crossover frequency: or, by simplifying and using the typical v ref = 1.25v: r c = 468k ? /v v out c out r dsp f c 4) determine the compensation capacitor to set the proper error-amplifier pole and zero determined from the above equations: core converter compensating the core converter is similar to the com- pensation of the main converter described above. the only difference is that the current is measured internal- ly, and the gain (transresistance) of the current-sense amplifier is r csc = 1.0v/a. the dc loop gain is: a vdc = 2000 v ref r load / v out c p co c 1 2r = r mf ap c vdc o c = ? 20 p c load max out out o i 2v = () p c load min out o 1 2r = () f c = f sw 5 fz 1 6c e co == /3 out sr frequency a vdc gain (db) phase 180 90 0 o phase phase margin z c = p o p c z 0 gain figure 6. current-mode step-down converter bode plot
max1802 digital camera step-down power supply ______________________________________________________________________________________ 23 to achieve a stable circuit for the core converter, use the following procedure: 1) determine the desired crossover frequency, either 1/3 of the zero due to the output capacitor esr: or 1/5 of the switching frequency: whichever is lower. 2) determine the pole frequency due to the output capacitor and the load resistor: or 3) determine the compensation resistor required to set the desired crossover frequency: or, by simplifying and using the typical v ref = 1.25v: r c = 50k ? /v v out c out f c 4) determine the compensation capacitor to set the proper error-amplifier pole and zero determined from the above equations: auxiliary controllers the auxiliary controllers use voltage mode to regulate their output voltages. the following explains how to compensate the control system for optimal perfor- mance. the compensation differs depending on whether the inductor current is continuous or discontin- uous. discontinuous inductor current for discontinuous inductor current, the pwm controller has a single pole. the pole frequency and dc gain of the pwm controller are dependent on the operating duty cycle, which is: d = (2 l f osc / r e ) 1/2 where r e is the equivalent load resistance, or: r e = v in 2 r load / (v out (v out - v in )) the frequency of single pole due to the pwm converter is: p o = (2 v out - v in ) / (2 (v out - v in ) r load c out ) and the dc gain of the pwm controller is: a vo = 2 v out (v out - v in ) r load / ((2 v out - v in ) d) note that, as in the current-mode, step-down cases above, as r load is increased, the pole frequency decreases and the dc gain increases proportionally. since the crossover frequency is the product of the pole frequency and the dc gain, it remains indepen- dent of the load. as in the cases of the main and core converters, the gain through the voltage-divider is a vdv = v ref / v out , and the dc gain of the error amplifier is a vea = 2000v/v. thus, the dc loop gain is a vdc = a vdv a vea a vo . the compensation resistor-capacitor pair at comp cause a pole and zero at frequencies (in hz): p c = g ea / (4000 c c ) = 1 / (4x10 7 c c ) z c = 1 / (2 r c c c ) and the esr of the output filter capacitor causes a zero in the loop response at the frequency (in hz): z o = 1 / (2 c out esr). the dc gain and the poles and zeros are shown in the bode plot of figure 7. to achieve a stable circuit with the bode plot of figure 7, follow the procedure below: 1) choose the r c that is equivalent to the inverse of the transconductance of the error amplifier, 1 / r c = g ea = 100s, or r c = 10k ? . this sets the high-fre- quency voltage gain of the error amplifier to 0db. 2) determine the maximum output pole frequency: where r load(min) = v out / i out(max) . p vc out load min out o(max) out in in 2v v 2vr = ? ? () () c p co c 1 2r = r mf ap c vdc o c = ? 20 p c load max out out o i 2v = () p c load min out o 1 2r = () f c = f sw 5 f z1 6c e c o == 3 out sr
max1802 digital camera step-down power supply 24 ______________________________________________________________________________________ 3) place the compensation zero at the same frequency as the maximum output pole frequency (in hz): solving for cc: use values of c c <10nf. if the above calculation deter- mines that the capacitor should be >10nf, use c c = 10nf, skip step 4, and go to step 5. 4) determine the crossover frequency (in hz): and to maintain at least 10db gain margin, make sure that the crossover frequency is 1/3 of the esr zero frequency, or 3f c z o , or esr d / 6 v ref . if this is not the case, go to step 5 to reduce the error- amplifier high-frequency gain to decrease the crossover frequency. 5) the high-frequency gain may be reduced, thus reducing the crossover frequency, as long as the zero due to the compensation network remains at or below the crossover frequency. in this case: and choose c out , r c , and c c to satisfy both equations simultaneously. continuous inductor current for continuous inductor current, there are two condi- tions that change, requiring different compensation. the response of the control loop includes a right-half- plane zero and a complex pole pair due to the inductor and output capacitor. for stable operation, the con- troller-loop gain must drop below unity (0db) at a much lower frequency than the right-half-plane zero frequen- cy. the zero arising from the esr of the output capaci- tor is typically used to compensate the control circuit by increasing the phase near the crossover frequency, increasing the phase margin. if a low-value, low-esr output capacitor (such as a ceramic capacitor) is used, the esr-related zero occurs at too high a frequency and does not increase the phase margin. in this case, use a lower value inductor so that it operates with dis- continuous current (see the discontinuous inductor current section). for continuous inductor current, the gain of the voltage divider is a vdv = v ref / v out, and the dc gain of the error amplifier is a vea = 2000. the gain through the pwm controller in continuous current is: thus, the total dc loop gain is: a vdc = 2000 v out / v in . the complex pole pair due to the inductor and output capacitor occurs at the frequency (in hz): the pole and zero due to the compensation network at comp occur at the frequencies (in hz): z 1 2r c c c c = p g c c c ea c c = () = 4000 1 410 7 p v vlc o out in out = 2 a vo out in ref v vv = 2 f rv c c ref = g drc ea c out c c 1 2 esr d g ea c rv ref 6 f v c c ref = d out cc v out out c out max c out in out in vv r i 2v v = ? ? () ? ? ? ? ? ? ? ? () z rc v c c c out load min out c out in in 2v v 2vr == ? ? () 1 2 () frequency a vdc gain (db) phase 180 90 0 o -20 20 40 60 80 phase z c = p o z 0 p c gain figure 7. discontinuous-current, voltage-mode, step-up controller bode plot
max1802 digital camera step-down power supply ______________________________________________________________________________________ 25 the frequency (in hz) of the zero due to the esr of the output capacitor is: and the right-half-plane zero frequency (in hz) is: figure 8 shows the bode plot of the loop gain of this control circuit. to configure the compensation network for a stable control loop, set the crossover frequency at that of the zero due to the output capacitor esr. use the following procedure: 1) determine the frequency of the right-half-plane zero: 2) find the dc loop gain: 3) determine the frequency of the complex pole pair due to the inductor and output capacitor: 4) since response is 2nd order (-40db per decade) between the complex pole pair and the esr zero, determine the desired amplitude at the complex pole pair to force the crossover frequency equal to the esr zero frequency. thus: 5) determine the desired compensation pole. since the response between the compensation pole and the complex pole pair is 1st order (-20db per decade), the ratio of the frequencies is equal to the ratio of the amplitudes at those frequencies. thus: solving this equation for c c : 6) determine r c for the compensation zero frequency as equal to the complex pole-pair frequency: z c = p o . solving for r c : applications information using the max1801 with the max1802 step-down master the max1801 is a slave dc-dc controller that can be used with the max1802 to generate additional output voltages. the max1801 does not generate its own ref- erence or oscillator. instead it uses the reference and oscillator from the max1802 step-down master convert- er controller (figure 1). max1801 controller operation and design is similar to that of the max1802 auxiliary controllers. for more details, refer to the max1801 data sheet. using an auxiliary controller in an sepic configuration where the battery voltage may be above or below the required output voltage, neither a step-up nor a step- down converter is suitable; instead, use a step-up/step- down converter. one type of step-up/step-down r v vc c in out c = lc out c vc e v c out out in = () ? () 32 2 12 20 / / sr ml p p a o c dc = () ap o ap v c o in () = () = zp l esr v oo out out / 2 2 2 2 f v o out = 2 vlc in out a v vdc in = 2000v out z 1-d 2 rhp = () 2 r l load z 1-d 2 rhp = () 2 r l load z 1 2c e o out = sr frequency a vdc phase 180 90 0 0 phase gain z c = p o z 0 z rhp phase margin gain margin a vdc gain (db) -10 10 20 30 p c 40 figure 8. continuous-current, voltage-mode, step-up converter bode plot
max1802 digital camera step-down power supply 26 ______________________________________________________________________________________ converter is the sepic, shown in figure 9. inductors l1 and l2 can be separate inductors or can be wound on a single core and coupled like a transformer. typically, using a coupled inductor will improve efficiency since some power is transferred through the coupling, so less power passes through the coupling capacitor (c2). likewise, c2 should have low esr to improve efficien- cy. the ripple current rating must be greater than the larger of the input and output currents. the mosfet (q1) drain-source voltage rating and the rectifier (d1) reverse-voltage rating must exceed the sum of the input and output voltages. other types of step-up/step- down circuits are a flyback converter and a step-up converter followed by a linear regulator. using an auxiliary controller for a multi-output flyback circuit some applications require multiple voltages from a sin- gle converter that features a flyback transformer. figure 10 shows a max1802 auxiliary controller in a two-output flyback configuration. the controller drives an external mosfet that switches the transformer pri- mary, and the two secondaries generate the outputs. only a single positive output voltage can be regulated using the feedback resistive voltage-divider, so the other voltages are set by the turns ratio of the trans- former secondaries. the regulation of the other sec- ondary voltages degrades due to transformer leakage inductance and winding resistance. voltage regulation is best when the load current is limited to a small range. consult the transformer manufacturer for the proper design for a given application. using a charge pump for negative output voltages negative output voltages can be produced without a transformer using a charge-pump circuit with an auxil- iary controller as shown in figure 11. when mosfet q1 turns off, the voltage at its drain rises to supply cur- rent to v out+ . at the same time, c1 charges to the volt- age at v out+ through d1. when the mosfet turns on, c1 discharges through d3, thereby charging c3 to v out- minus the drop across d3 to create roughly the same voltage as v out+ at v out- but with inverted polarity. if different magnitudes are required for the positive and negative voltages, a linear regulator can be used at one of the outputs to achieve the desired voltage. designing a pc board a good pc board layout is important to achieve optimal performance from the max1802. good design reduces excessive conducted and/or radiated noise, both of which are undesirable. conductors carrying discontinuous currents should be kept as short as possible. conductors carrying high currents should be made as wide as possible. a sepa- rate low-noise ground plane containing the reference and signal grounds should only connect to the power- ground plane at one point to minimize the effects of power-ground currents. keep the voltage feedback network very close to the ic, preferably within 0.2in (5mm) of the fb_ pin. nodes with high dv/dt (switching nodes) should be kept as small as possible and should stay away from high- impedance nodes such as fb_ and comp_. refer to the max1802evkit evaluation kit manual for a full pc board example. chip information transistor count: 7740 r c g c max1802 d1 l2 l1 c2 r1 output 3.3v r2 q1 input 1 cell li+ main on comp dcon ext fb figure 9. auxiliary controller, sepic configuration
max1802 digital camera step-down power supply ______________________________________________________________________________________ 27 r c g c max1802 + output - output r1 r2 q1 input 1 cell li+ main on comp dcon ext fb figure 10. auxiliary controller, flyback configuration r c g c max1802 v out - v out + d3 d2 2 d1 c3 c2 r1 r2 c1 q1 l input 1 cell li+ main on comp dcon ext fb figure 11. auxiliary controller, charge-pump configuration pin configuration max1802 tqfp top view 32 28 29 30 31 25 26 27 ref fbc compc vl gnd vddc lxc pgnd 10 13 15 14 16 11 12 9 pgndm dcon1 osc on1 dl1 fb1 comp1 fb2 17 18 19 20 21 22 23 dcon3 24 onc fb3 comp3 dl3 dl2 dcon2 comp2 2 3 4 5 6 7 8 dlm lxm dhm vddm vh onm compm 1 fbm
max1802 digital camera step-down power supply maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a maxim product. no circu it patent licenses are implied. maxim reserves the right to change the circuitry and specifications without notice at any time. 28 ____________________maxim integrated products, 120 san gabriel drive, sunnyvale, ca 94086 408-737-7600 ? 2000 maxim integrated products printed usa is a registered trademark of maxim integrated products. maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a maxim product. no circu it patent licenses are implied. maxim reserves the right to change the circuitry and specifications without notice at any time. 28 ____________________maxim integrated products, 120 san gabriel drive, sunnyvale, ca 94086 408-737-7600 ? 2000 maxim integrated products printed usa is a registered trademark of maxim integrated products. 32l tqfp, 5x5x01.0.eps package information

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