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fn8566 rev 1.00 page 1 of 35 november 2, 2015 fn8566 rev 1.00 november 2, 2015 isl95712 multiphase pwm regulator for amd fusion? desktop cpus using svi 2.0 datasheet the isl95712 is fully compliant with amd fusion? svi 2.0 and provides a complete solution for microprocessor and graphics processor core power. the isl95712 controller supports two voltage regulators (vrs) for core and northbridge outputs. the core vr can be configured for 4-, 3-, 2-, or 1-phase operation while the northbridge vr supports 3-, 2- or 1-phase configurations for maximum flexibility. the two vrs share a serial control bus to communicate with the amd cpu and achieve lower cost and smaller board area compared with two-chip solutions. the pwm modulator is based on intersil?s robust ripple regulator r3? technology. compar ed to traditional modulators, the r3? modulator can automatically change switching frequency for faster transient sett ling time during load transients and improved light load efficiency. the isl95712 has several other key features. both outputs support dcr current sensing with a single ntc thermistor for dcr temperature compensation or accurate resistor current sensing. they also utilize remote voltage sense, adjustable switching frequency, oc protection and power-good indicators. applications ? amd fusion cpu/gpu core power ?desktop computers features ? supports amd svi 2.0 serial data bus interface and pmbus - serial vid clock frequency range 100khz to 25mhz ? dual output controller with 12v integrated core gate drivers ? precision voltage regulation - 0.5% system accuracy over-temperature - 0.5v to 1.55v in 6.25mv steps - enhanced load line accuracy ? supports multiple current sensing methods - lossless inductor dcr current sensing - precision resistor current sensing ? programmable 1-, 2-, 3- or 4-phase for the core output and 1- , 2- or 3-phase for the northbridge output ? adaptive body diode conduction time reduction ? superior noise immunity and transient response ? output current and voltage telemetry ? differential remote voltage sensing ? high efficiency across entire load range ?programmable slew rate ? programmable vid offset and droop on both outputs ? programmable switching frequency for both outputs ? excellent dynamic current balance between phases ? protection: ocp/woc, ovp, pgood and thermal monitor ? small footprint 52 ld 6x6 qfn package - pb-free (rohs compliant) performance figure 1. efficiency vs load figure 2. v out vs load 0 10 20 30 40 50 60 70 80 90 100 0 10 20 30 40 50 60 70 80 90 100 110 efficiency (%) load current (a) dac = 1.500v northbridge core core (psi1) 1.0 1.1 1.2 1.3 1.4 1.5 1.6 0 10 20 30 40 50 60 70 80 90 100 110 output voltage (v) load current (a) dac = 1.500v northbridge core
isl95712 fn8566 rev 1.00 page 2 of 35 november 2, 2015 table of contents simplified application circuit for high power cpu core . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 pin configuration. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 pin descriptions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 thermal information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 recommended operating conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 electrical specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 gate driver timing diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 theory of operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 multiphase r3? modulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 diode emulation and period stretching . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 channel configuration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 power-on reset . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11 start-up timing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 diode throttling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 voltage regulation and load line implementation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 differential sensing. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 phase current balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 modes of operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15 dynamic operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 adaptive body diode conduction time reduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 resistor configuration options. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 vr offset programming . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 vid-on-the-fly slew rate selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 ccm switching frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 amd serial vid interface 2.0 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 pre-pwrok metal vid. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 svi interface active . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 vid-on-the-fly transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 svi data communication protocol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 svi bus protocol. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 power states . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 dynamic load line slope trim . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 dynamic offset trim . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 telemetry. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 pmbus interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 protection features. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 overcurrent . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 current-balance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24 undervoltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 overvoltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 thermal monitor [ntc, ntc_nb] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25 fault recovery . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 interface pin protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 key component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 inductor dcr current-sensing network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 resistor current-sensing network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 overcurrent protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 load line slope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 compensator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 current balancing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 thermal monitor component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 layout guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 pcb layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 revision history. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 about intersil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 package outline drawing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
isl95712 fn8566 rev 1.00 page 3 of 35 november 2, 2015 simplified application circui t for high power cpu core nb_ph1 nb_ph2 figure 3. typical application circuit using inductor dcr sensing boot_nb ugate_nb phase_nb lgate_nb boot2 ugate2 phase2 lgate2 boot1 ugate1 phase1 lgate1 pwm3 isump isumn ph1 ph2 ph3 vo1 vo2 vo3 gnd pad fb_nb comp_nb vsen_nb vnb_sense vddp pgood isen1 isen2 isen3 enable ph1 ph2 ph3 vcore ph2 vo2 vo1 +12v vo3 ph1 +12v +12v vo1 ph3 ph1 +12v vnb vdd pwm2_nb isl6625a vnb1 nb_ph1 vnb2 nb_ph2 isen1_nb isen2_nb isump_nb isumn_nb nb_ph1 nb_ph2 vnb1 vnb2 ntc_nb imon imon_nb pwrok svt svd p svc vddio ntc vr_hot_l thermal indicator prog vcore_sense fb comp vsen rtn +12v ntc ntc cn cn ri ri *optional * * *optional * * isen4 ph4 vo4 ph4 pwm4 +12v vo4 ph4 pgood_nb isl95712 i2data i2clk isl6625a isl6625a
isl95712 fn8566 rev 1.00 page 4 of 35 november 2, 2015 pin configuration isl95712 (52 ld qfn) top view 1 52 gnd (bottom pad) c o m p _ n b i s u m p _ n b 2 3 ntc_nb 4 svc 5 6 vr_hot_l 7 imon 8 vddio 9 svt 10 enable 11 svd 12 ntc 51 50 49 v s e n _ n b 48 p w m 3 _ n b 47 p w m 2 _ n b 46 45 44 i m o n _ n b 43 42 41 p w m 4 39 pwm3 38 37 phase1_nb 36 ugate2 35 lgate1_nb 34 lgate2 33 vddp 32 31 30 ugate1_nb 29 phase2 28 boot1_nb 14 b o o t 1 15 v s e n 16 i s u m n 17 v d d 18 19 f b 20 21 22 i s e n 4 23 24 r t n p h a s e 1 pwrok i s u m p f b _ n b i s u m n _ n b i s e n 1 _ n b lgate1 i i s e n 2 i s e n 1 i s e n 3 boot2 p r o g p g o o d _ n b i 2 d a t a i 2 c l k p g o o d 40 27 ugate1 13 25 26 isen3_nb c o m p i s e n 2 _ n b pin descriptions pin number symbol description 1 isen3_nb individual current sensing for channel 3 of the northbridge vr. when isen3_nb is pulled to +5v, the controller will disable channel 3 and the northbridge vr will run 2-phase. 2 ntc_nb thermistor input to vr_hot_l circuit to monitor northbridge vr temperature. 3 imon_nb northbridge output cu rrent monitor. a current proportional to the northbridge vr output current is sourced from this pin. 4svc serial vid clock input from the cpu processor master device. 5 vr_hot_l thermal indicator signal to amd cpu. therma l overload open-drain output indicator active low. 6 svd serial vid data bidirectional signal from the cpu processor master device to the vr. 7 vddio vddio is the processor memory interface power rail an d this pin serves as the reference to the controller ic for this processor i/o signal level. 8 svt serial vid telemetry (svt) data line input to the cpu from the controller ic. telemetry and vid-on-the-fly complete signal provided from this pin. 9 enable enable input. a high level lo gic on this pin enables both vrs. 10 pwrok system power-good input. when this pin is high, the svi 2 interface is active and the i 2 c protocol is running. while this pin is low, the svc and svd in put states determine the pre-pwrok metal vid. this pin must be low prior to the isl95712 pgood output going high per the amd svi 2.0 controller guidelines. 11 imon core output current monitor. a current proportional to the core vr output current is sourced from this pin. 12 ntc thermistor input to vr_hot_l circuit to monitor core vr temperature. 13 isen4 isen4 is the individual current sensing for channel 4 of the core vr. when isen4 is pulled to +5v, the controller disables channel 4, and the core vr runs in three-phase mode.
isl95712 fn8566 rev 1.00 page 5 of 35 november 2, 2015 14 isen3 isen3 is the individual current sensing for channe l 3 of the core vr. when isen3 is pulled to +5v, the controller disables channel 3, and the core vr runs in two-phase mode. 15 isen2 individual current sensing for channel 2 of the core vr. when isen2 is pulled to +5v, the controller disables channel 2, and the core vr runs in single-phase mode. 16 isen1 individual current sensing for channel 1 of the core vr. if isen2 is tied to +5v , this pin cannot be left open and must be tied to gnd with a 10k resistor. if isen1 is tied to +5v , the core portion of the ic is shut down. 17 isump noninverting input of the tran sconductance amplifier for current monitor and load line of core output. 18 isumn inverting input of the transconductance amplifie r for current monitor and load line of core output. 19 vsen output voltage sense pin for the core controller. connect to the +sense pin of the microprocessor die. 20 rtn output voltage sense return pin for both core vr and northbridge vr . connect to the -sense pin of the microprocessor die. 21 fb output voltage feedback to the inverting input of the core controller error amplifier. 22 vdd 5v bias power. a resistor [2 ] and a decoupling capacitor should be used from the +5v supply. a high quality, x7r dielectric mlcc capacitor is recommended. 23 pgood open-drain output to indicate the core output is ready to supply regulated voltage. pull-up externally to vdd or 3.3v through a resistor. 24 comp core controller error amplifier output. a resistor from comp to gnd sets the core vr offset voltage. 25 boot1 connect an mlcc capacitor across the boot1 and phase1 pins. the boot capacitor is charged, through an internal boot diode connected from the vddp pin to the boot1 pin, each time the phase1 pin drops below vddp minus the voltage dropped across the internal boot diode. 26 phase1 current return path for the phase 1 high-side mosf et gate driver of vr1. connect the phase1 pin to the node consisting of the high-side mo sfet source, the low-side mosfet drain and the output inductor of phase 1. 27 ugate1 output of the phase 1 high-side mosfet gate driv er of the core vr. connect the ugate1 pin to the gate of the phase 1 high-side mosfet(s). 28 lgate1 output of the phase 1 low-side mosfet gate driv er of the core vr. connect the lgate1 pin to the gate of the phase 1 low-side mosfet(s). 29 boot2 connect an mlcc capacitor across the boot2 and phase2 pins. the boot capacitor is charged, through an internal boot diode connected from the vddp pin to the boot2 pin, each time the phase2 pin drops below vddp minus the voltage dropped across the internal boot diode. 30 phase2 current return path for the phase 2 high-side mo sfet gate driver of the core vr. connect the phase2 pin to the node consisting of th e high-side mosfet source, the low- side mosfet drain and the output inductor of phase 2. 31 ugate2 output of the phase 2 high-side mosfet gate driv er of the core vr. connect the ugate2 pin to the gate of the phase 2 high-side mosfet(s). 32 vddp input voltage bias for the internal gate drivers. co nnect +12v to the vddp pin. decouple with at least 1f of capacitance to gnd. a high quality, x7r dielectric mlcc capacitor is recommended. 33 lgate2 output of the phase 2 low-side mosfet gate driv er of the core vr. connect the lgate2 pin to the gate of the phase 2 low-side mosfet(s). 34 lgate1_nb output of northbridge phase 1 low-side mosfet gate driver. connect the lgate1_nb pin to the gate of the northbridge vr phase 1 low-side mosfet(s). 35 phase1_nb current return path for northbridge vr phas e 1 high-side mosfet gate driver. connect the phase1_nb pin to the node consisting of th e high-side mosfet source, the low- side mosfet drain and the output inductor of northbridge phase 1. 36 ugate1_nb output of the phase 1 high-side mosfet gate driver of the northbridge vr. connect the ugate1_nb pin to the gate of the northbridge vr phase 1 high-side mosfet(s). pin descriptions (continued) pin number symbol description
isl95712 fn8566 rev 1.00 page 6 of 35 november 2, 2015 37 boot1_nb connect an mlcc capacitor across the boot1_nb and phase1_nb pins. the boot capacitor is charged, through an internal boot diode connected from th e vddp pin to the boot1_nb pin, each time the phase1_nb pin drops below vddp minus the voltage dropped across the internal boot diode. 38 pwm3 pwm output of channel 3 of the core vr. disabled if isen3 is tied to +5v. 39 pwm4 pwm output of channel 4 of the core vr. disabled if isen4 is tied to +5v. 40 pwm2_nb pwm output for channel 2 of the northbri dge vr. disabled when isen2_nb is tied to +5v. 41 pwm3_nb pwm output for channel 3 of the northbri dge vr. disabled when isen3_nb is tied to +5v. 42, 43 i2clk, i2data smbus/pmbus/i 2 c interface used for additional communication with the controller outside of the svi2 pins. tie to vcc with 4.7k pull-up resistor when not used. 44 prog a resistor from the prog pin to gnd programs the switching frequency. 45 pgood_nb open-drain output to indicate the northbridge output is ready to supply regulated voltage. pull-up externally to vdd or 3.3v through a resistor. 46 comp_nb northbridge vr error amplifie r output. a resistor from comp_nb to gnd sets the northbridge vr offset voltage and is used to set the switching fr equency for the core vr and northbridge vr. 47 fb_nb output voltage feedback to the inverting in put of the northbridge controller error amplifier. 48 vsen_nb output voltage sense pin for the northbridge cont roller. connect to the +sense pin of the microprocessor die. 49 isumn_nb inverting input of the transconductance amplifier for current monitor and load line of the northbridge vr. 50 isump_nb noninverting input of the transconductance amplif ier for current monitor and load line of the northbridge vr. 51 isen1_nb individual current sensing for channel 1 of the nort hbridge vr. if isen1_nb is tied to +5v, this pin cannot be left open and must be tied to gnd with a 10k resistor. if isen1_nb is tied to +5v, the northbridge portion of the ic is shutdown. 52 isen2_nb individual current sensing for channel 2 of the northbridge vr. when isen2_nb is pulled to +5v, the controller will disable channels 2 and 3 and the northbridge vr will run 1-phase. gnd (bottom pad) signal common of the ic. unless otherwise stated, signals are referenced to the gnd pin. pin descriptions (continued) pin number symbol description ordering information part number (notes 1, 2, 3) part marking temp. range (c) package (rohs compliant) pkg. dwg. # isl95712hrz 95712 hrz -10 to +100 52 ld 6x6 qfn l52.6x6a ISL95712IRZ 95712 irz -40 to +100 52 ld 6x6 qfn l52.6x6a notes: 1. add ?-t? suffix for tape and reel. please refer to tb347 for details on reel specifications. 2. intersil pb-free plus anneal products employ special pb-free material sets; molding compounds/die attach materials and 100% m atte tin plate termination finish, which are rohs compliant and compatible with both snpb and pb-free soldering operations. intersil pb-free p roducts are msl classified at pb-free peak reflow temp eratures that meet or exceed the pb-fr ee requirements of ipc/jedec j std-020. 3. for moisture sensitivity level (msl), please see device information page for isl95712 . for more information on msl please see tech brief tb363 .
isl95712 fn8566 rev 1.00 page 7 of 35 november 2, 2015 absolute maximum rating s thermal information supply voltage, v dd . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3v to +7v input supply voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +15v gate driver supply voltage, v ddp . . . . . . . . . . . . . . . . . . . . . . -0.3v to + 15v boot voltage (v boot ) . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3v to v ddp + 15v ugate voltage (v ugate ). . . . . . . . . . . . . . v phase - 0.3v dc to v boot + 0.3v v phase - 3.5v (<100ns pulse width, 2j) to v boot + 0.3v lgate voltage (v lgate ) . . . . . . . . . . . . . . . . . gnd - 0.3v dc to v ddp + 0.3v gnd - 5v (<100ns pulse width, 2j) to v ddp + 0.3v phase voltage (v phase ) . . . . . . . . . . . . . . . . . . . . . gnd - 0.3v dc to 25v dc gnd - 8v (>400ns pulse width, 20) to 30v (<200ns) open-drain outputs, pgood, pgood_nb, vr_hot_l. . . . . . . -0.3v to +7v all other pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-0.3v to vdd + 0.3v thermal resistance (typical) ? ja (c/w) ? jc (c/w) 52 ld qfn package ( notes 4 , 5 ) . . . . . . . . 28 2.5 maximum junction temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . .+150c maximum storage temperature range . . . . . . . . . . . . . .-65c to +150c pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see tb493 recommended operating conditions supply voltage, v dd . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +5v 5% input supply and gate drive voltages, v ddp . . . . . . . . . . . . . . . +12v 5% ambient temperature hrz. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10c to +100c irz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40c to +100c junction temperature hrz. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-10c to +125c irz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40c to +125c caution: do not operate at or near the maximum ratings listed for extended periods of time. exposure to such conditions may adv ersely impact product reliability and result in failures not covered by warranty. notes: 4. ? ja is measured in free air with the componen t mounted on a high effective thermal conduc tivity test board with ?direct attach? fe atures. see tech brief tb379 . 5. for ? jc , the ?case temp? location is the center of the exposed metal pad on the package underside. electrical specifications operating conditions: v dd = 5v, t a = -10c to +100c (hrz), f sw = 300khz, unless otherwise noted. boldface limits apply across the operating temperature range, -40c to +100c. parameter symbol test conditions min ( note 6 )typ max ( note 6 )unit input power supply +5v supply current i vdd enable = 1v 12.5 14.0 ma enable = 0v 125 a power-on-reset thresholds vdd por threshold vdd_por r v dd rising 4.35 4.50 v vdd_por f v dd falling 4.00 4.15 v system and references system accuracy hrz %error (v out) no load; closed loop, active mode range, vid = 0.75v to 1.55v -0.5 +0.5 % vid = 0.25v to 0.74375v -10 +10 mv irz %error (v out ) no load; closed loop, active mode range, vid = 0.75v to 1.55v -0.8 +0.8 % vid = 0.25v to 0.74375v -12 +12 mv maximum output voltage v out(max) vid = [00000000] 1.55 v minimum output voltage v out(min) vid = [11111111] 0 v channel frequency nominal channel frequency f sw(nom) 280 300 320 khz adjustment range 300 450 khz amplifiers current-sense amplifier input offset hrz i fb = 0a -0.15 +0.15 mv irz i fb = 0a -0.20 +0.20 mv error amp dc gain a v0 119 db error amp gain-bandwidth product gbw c l = 20pf 17 mhz isen input bias current 20 na
isl95712 fn8566 rev 1.00 page 8 of 35 november 2, 2015 power-good (pgood and pgood_nb) and protection monitors pgood low voltage v ol i pgood = 4ma 0.4 v pgood leakage current i oh pgood = 3.3v -1 1 a pwrok high threshold 750 mv vr_hot_l pull-down 11 pwrok leakage current 1 a vr_hot_l leakage current 1 a gate driver ugate pull-up resistance r ugpu 200ma source current 1.0 1.5 ugate source current i ugsrc ugate - phase = 2.5v 2 a ugate sink resistance r ugpd 250ma sink current 1.0 1.5 ugate sink current i ugsnk ugate - phase = 2.5v 2 a lgate pull-up resistance r lgpu 250ma source current 1.0 1.5 lgate source current i lgsrc lgate - vssp = 2.5v 2 a lgate sink resistance r lgpd 250ma sink current 0.5 0.9 lgate sink current i lgsnk lgate - vssp = 2.5v 4 a ugate to lgate dead time t ugflgr ugate falling to lgate rising, no load 59 ns lgate to ugate dead time t lgfugr lgate falling to ugate rising, no load 37 ns protection overvoltage threshold ov th vsen rising above setpoint for >1s 275 325 375 mv undervoltage threshold uv th vsen falls below setpoint for >1s 275 325 375 mv current imbalance threshold one isen above another isen for >1.2ms 9 mv way overcurrent trip threshold [imonx current based detection] imonx woc all states, i droop = 60a, r imon = 135k 15 a overcurrent trip threshold [imonx voltage based detection] v imonx_ocp all states, i droop = 45a, i imonx = 11.25a, r imon = 135k 1.485 1.510 1.535 v logic thresholds enable input low v il 1 v enable input high v ih hrz 1.6 v v ih irz 1.65 v enable leakage current i enable enable = 0v -1 01a enable = 1v 1 a svt impedance 50 svc, svd input low v il % of vddio 30 % svc, svd input high v ih % of vddio 70 % svc, svd leakage enable = 0v, svc, svd = 0v and 1v -1 1 a enable = 1v, svc, svd = 1v -5 1 a enable = 1v, svc, svd = 0v -35 -20 -5 a pwm pwm output low v 0l sinking 5ma 1 v pwm output high v 0h sourcing 5ma 3.5 v pwm tri-state leakage pwm = 2.5v 0.5 a thermal monitor ntc source current ntc = 0.6v 27 30 33 a ntc thermal warning voltage 600 640 680 mv electrical specifications operating conditions: v dd = 5v, t a = -10c to +100c (hrz), f sw = 300khz, unless otherwise noted. boldface limits apply across the operating temp erature range, -40c to +100c. (continued) parameter symbol test conditions min ( note 6 )typ max ( note 6 )unit
isl95712 fn8566 rev 1.00 page 9 of 35 november 2, 2015 gate driver timing diagram ntc thermal warning voltage hysteresis 20 mv ntc thermal shutdown voltage 530 580 630 mv slew rate vid-on-the-fly slew rate maximum programmed 16 20 24 mv/s minimum programmed 8 10 12 mv/s note: 6. compliance to datasheet limits is assu red by one or more methods: production test, characterization and/or design. electrical specifications operating conditions: v dd = 5v, t a = -10c to +100c (hrz), f sw = 300khz, unless otherwise noted. boldface limits apply across the operating temp erature range, -40c to +100c. (continued) parameter symbol test conditions min ( note 6 )typ max ( note 6 )unit pwm ugate lgate 1v 1v t ugflgr t rl t fu t ru t fl t lgfugr figure 4. gate driver timing diagram
isl95712 fn8566 rev 1.00 page 10 of 35 november 2, 2015 theory of operation multiphase r3? modulator the isl95712 is a multiphase regulator implementing two voltage regulators, core vr and northbridge (nb) vr, on one chip controlled by amd?s? svi2? protocol. the core vr can be programmed for 1-, 2-, 3- or 4-ph ase operation. the northbridge vr can be configured for 1-, 2-, or 3-phase operation. both regulators use the intersil patented r3? (robust ripple regulator) modulator. the r3? modulator combines the best features of fixed frequency pwm and hysteretic pwm while eliminating many of their shortcomings. figure 5 conceptually shows the multiphase r3? modulator circuit, and figure 6 shows the operation principles. inside the ic, the modulator uses the master clock circuit to generate the clocks for the slave circuits. the modulator discharges the ripple capacitor c rm with a current source equal to g m v o , where g m is a gain factor. c rm voltage v crm is a sawtooth waveform traversing between the vw and comp voltages. it resets to vw when it hits comp, and generates a one-shot master clock signal. a phase sequencer distributes the master clock signal to the slave circuits. if the core vr is in 4-phase mode, the master clock signal is distributed to the four phases, and the clock 1~4 signals wi ll be 90 out-of-phase. if the core vr is in 3-phase mode, the master clock signal is distributed to the three phases, and the clock 1~3 signals will be 120 out-of-phase. if the core vr is in 2-phase mode, the master clock signal is distributed to phases 1 and 2, and the clock1 and clock2 signals will be 180 out- of-phase. if the core vr is in 1-phase mode, the master clock signal will be distributed to phase 1 only and will be the clock1 signal. each slave circuit has its own ripple capacitor c rs , whose voltage mimics the inductor ripple current. a g m amplifier converts the inductor voltage into a current source to charge and discharge c rs . the slave circuit turns on it s pwm pulse upon receiving the clock signal, and the current source charges c rs . when c rs voltage v crs hits vw, the slave circuit turns off the pwm pulse, and the current source discharges c rs . since the controller works with v crs , which are large amplitude and noise-free synthesized signals, it achieves lower phase jitter than conventional hysteretic mode and fixed pwm mode controllers. unlike conventional hysteretic mode converters, the error amplifier allows the isl95712 to maintain a 0.5% output voltage accuracy. figure 7 shows the operation principles during load insertion response. the comp voltage ri ses during load insertion, generating the master clock sign al more quickly, so the pwm pulses turn on earlier, increasing the effective switching frequency. this allows for higher control loop bandwidth than conventional fixed frequency pwm controllers. the vw voltage rises as the comp voltage rises, making the pwm pulses wider. during load release response, th e comp voltage falls. it takes the master clock circuit longer to generate the next master clock signal so the pwm pulse is held off until needed. the vw voltage falls as the comp voltage falls, reducing the current pwm pulse width. this kind of behavior gives the isl95712 excellent response speed. the fact that all the phases share the same vw window voltage also ensures excellent dynamic current balance among phases. figure 5. r3 ? modulator circuit crm gmvo master clock vw comp master clock phase sequencer clock1 clock2 r i l1 gm clock1 phase1 crs1 vw s q pwm1 l1 r i l2 gm clock2 phase2 crs2 vw s q pwm2 l2 co vo vcrm vcrs1 vcrs2 master clock circuit slave circuit 1 slave circuit 2 r i l3 gm clock3 phase3 crs3 vw s q pwm3 l3 vcrs3 slave circuit 3 clock3 figure 6. r3 ? modulator operation principles in steady state comp vcrm master clock pwm1 vw clock1 pwm2 clock2 hysteretic window pwm3 vcrs3 clock3 vcrs2 vcrs1 vw
isl95712 fn8566 rev 1.00 page 11 of 35 november 2, 2015 diode emulation and period stretching the isl95712 can operate in di ode emulation (de) mode to improve light-load efficiency. in de mode, the low-side mosfet conducts when the current is flowing from source-to-drain and does not allow reverse curren t, thus emulating a diode. figure 8 shows when lgate is on, the low-side mosfet carries current, creating negative voltage on th e phase node due to the voltage drop across the on-resistance. th e isl95712 monitors the current by monitoring the phase node voltage. it turns off lgate when the phase node voltage reaches zero to prevent the inductor current from reversing the direction and creating unnecessary power loss. if the load current is light enough, as figure 8 shows, the inductor current reaches and stays at zero before the next phase node pulse, and the regulator is in discontinuous conduction mode (dcm). if the load current is heavy enough, the inductor current will never reach 0a, and the regulator is in ccm, although the controller is in de mode. figure 9 shows the operation principle in diode emulation mode at light load. the load gets incrementa lly lighter in each of the three cases from top to bottom. the pwm on-time is determined by the vw window size and therefore is the same, making the inductor current triangle the same in each of the three cases. the isl95712 clamps the ripple capacitor voltage v crs in de mode to make it mimic the inductor current. it takes the comp voltage longer to hit v crs , naturally stretching the sw itching period. the inductor current triangles move farther apart, such that the inductor current average value is equal to the load current. the reduced switching frequency helps increase light-load efficiency. channel configuration individual pwm channels of either vr can be disabled by connecting the isenx pin of the channel not required to +5v. for example, placing the controller in a 3+1 configuration, requires isen4 of the core vr and is en2_nb and isen3_nb of the northbridge vr to be tied to +5v. this disables channel 4 of the core vr and channels 2 and 3 of the northbridge vr. isen1_nb must be tied through a 10k resistor to gnd to prevent this pin from pulling high and disabling the channel. similarly, if the core vr is set to single phase mode, isen4, isen3 and isen2 will be tied to +5v while isen1 is tied to gnd through a 10k resistor. connecting isen1 or isen1_nb to +5v will disable the corresponding vr output. this feature allows debugging of individual vr outputs. power-on reset before the controller has suffic ient bias to guarantee proper operation, the isl95712 requires a +5v input supply tied to vdd to exceed the vdd rising power-on reset (por) threshold. once this threshold is reached or exceeded, the isl95712 has enough bias to check the state of the svi inputs once enable is taken high. hysteresis between the rising and the falling thresholds assure the isl95712 does not inadvertently turn off unless the bias voltage drops substantially (see ?electrical specifications? on page 7 ). note that v in must be present for the controller to drive the output voltage. figure 7. r3 ? modulator operation principles in load insertion response comp v crm master clock pwm1 vcrs1 vw clock1 pwm2 vcrs2 clock2 pwm3 clock3 vcrs3 vw pwm ugate phase il lgate figure 8. diode emulation il il v crs il v crs v crs vw ccm/dcm boundary light dcm deep dcm vw vw figure 9. period stretching
isl95712 fn8566 rev 1.00 page 12 of 35 november 2, 2015 start-up timing with vdd above the por threshold, the controller start-up sequence begins when enable ex ceeds the logic high threshold. figure 11 shows the typical soft-start timing of the core and northbridge vrs. once the controller registers enable as a high, the controller checks the state of a few programming pins during the typical 8ms delay prior to beginning soft-starting the core and northbridge outputs. the pre-pwrok metal vid is read from the state of the svc and svd pins and programs the dac, the programming resistors on the comp, comp_nb and prog pins are read to configure switching frequency, slew rate and output offsets. these programming resistors are discussed in subsequent sections. the isl95712 use a digital soft-start to ramp up the dac to the metal vid level programmed. the soft-start slew rate is programmed by the prog resistor, which is used to set the vid-on-the-fly slew rate as well. see the ? vid-on-the-fly slew rate selection ? on page 17 for more details on selecting the prog resistor. pgood is asserted high at the end of the soft-start ramp. diode throttling during the soft-start ramp-up, the isl95712 operates in diode throttling mode until the output has exceeded 400mv. in diode throttling mode, the lower mosfet is kept off so that the mosfet body diode conducts, similar to a standard buck regulator. voltage regulation and load line implementation after the soft-start sequence, th e isl95712 regulates the output voltages to the pre-pwrok metal vid programmed, see table 6 on page 17 . the isl95712 controls the no-load output voltage to an accuracy of 0.5% over the range of 0.75v to 1.55v. a differential amplifier allows voltage sensing for precise voltage regulation at the microprocessor die. vdd svc svd enable pwrok v core / v core_nb 1 7 8 figure 10. svi interface timing diagra m: typical pre-pwrok metal vid start-up pgood and pgood_nb 3 4 2 5 6 metal_vid v_svi interval 1 to 2: isl95712 waits to por. interval 2 to 3: svc and svd are externally set to pre-metal vi d code. interval 3 to 4: enable locks pre -metal vid code. both outputs soft-start to this level. interval 4 to 5: pgood signal goes high, indicating proper oper ation. interval 6 to 7: svc and svd da ta lines communicate change in v id code. interval 7 to 8: isl95712 responds to vid-on-the-fly code chang e and issues a votf for positive vid changes. interval 5 to 6: pgood and pgoo d_nb high is detected and pwrok is taken high. the isl95712 is prepared for svi commands. svt telemetry telemetry votf post 8: telemetry is clocked out of the isl95712. vdd enable dac 8ms metalvid slew rate vid command voltage pgood pwrok vin figure 11. typical soft-start waveforms
isl95712 fn8566 rev 1.00 page 13 of 35 november 2, 2015 as the load current increases from zero, the output voltage droops from the vid programmed value by an amount proportional to the load current, to achieve the load line. the isl95712 can sense the inductor current through the intrinsic dc resistance (dcr) of the inductors, as shown in figures 13 and 14 , or through resistors in series with the inductors, as shown in figure 25 on page 28 . in both methods, capacitor c n voltage represents the total inductor current. an internal amplifier converts c n voltage into an internal current source, i sum , with the gain set by resistor r i , see equation 1 . the i sum current is used for load line implementation, current monitoring on the imon pins and overcurrent protection. figure 12 shows the load line implementation. the isl95712 drives a current source (i droop ) out of the fb pin, which is a ratio of the i sum current, as described by equation 2 . when using inductor dcr current sensing, a single ntc element is used to compensate the positive temperature coefficient of the copper winding, thus sustaining the load line accuracy with reduced cost. i droop flows through resistor r droop and creates a voltage drop as shown in equation 3 . v droop is the droop voltage required to implement load line. changing r droop or scaling i droop can change the load line slope. since i sum sets the overcurrent protection level, it is recommended to first scale i sum based on ocp requirement, then select an appropriate r droop value to obtain the desired load line slope. differential sensing figure 12 also shows the differential voltage sensing scheme. vcc sense and vss sense are the remote voltage sensing signals from the processor die. a unity ga in differential amplifier senses the vss sense voltage and adds it to the dac output. the error amplifier regulates the inverting an d noninverting input voltages to be equal as shown in equation 4 : rewriting equation 4 and substituting equation 3 gives equation 5 the exact equation required fo r load line implementation. the vcc sense and vss sense signals come from the processor die. the feedback is open circuit in the absence of the processor. as figure 12 shows, it is recommended to add a ?catch? resistor to feed the vr local output voltage back to the compensator, and to add another ?catch? resistor to co nnect the vr local output ground to the rtn pin. these resistors, typically 10 , provide voltage feedback if the system is powered up without a processor installed. phase current balancing the isl95712 monitors individual phase average current by monitoring the isen1, isen2, isen3 and isen4 voltages. figure 13 shows the recommended current balancing circuit for dcr sensing. each phase node vo ltage is averaged by a low-pass filter consisting of r isen and c isen , and is presented to the corresponding isen pin. r isen should be routed to the inductor phase-node pad in order to eliminate the effect of phase node parasitic pcb dcr. equations 6 through 9 give the isen pin voltages: where r dcr1 , r dcr2 , r dcr3 and r dcr4 are inductor dcr; r pcb1 , r pcb2 , r pcb3 and r pcb4 are parasitic pcb dcr between the inductor output side pad and the output voltage rail; and i l1 , i l2 , i l3 and i l4 are inductor average currents. figure 12. differential sensing and load line implementation x 1 e/a ? dac svid[7:0] r droop i droop vdac v droop fb comp vcc sense vss sense rtn vss internal to ic ?catch? resistor ?catch? resistor vr local vo + - +- + + - svc svd i sum v cn r i ---------- - = (eq. 1) i droop 5 4 -- - i sum ? 5 4 -- - v cn r i ---------- - ? == (eq. 2) v droop r droop i droop ? = (eq. 3) vcc sense v + droop v dac vss sense + = (eq. 4) vcc sense vss sense C v dac r droop i droop ? C = (eq. 5) figure 13. current balancing circuit v o isen3 l3 r isen c isen isen2 r isen c isen isen1 r isen c isen l2 l1 r dcr3 r dcr2 r dcr1 phase3 phase2 phase1 i l3 i l2 i l1 r pcb3 r pcb2 r pcb1 l4 r dcr4 i l4 r pcb4 phase4 r isen c isen isen4 v isen1 r dcr1 r pcb1 + ?? i l1 ? = (eq. 6) v isen2 r dcr2 r pcb2 + ?? i l2 ? = (eq. 7) v isen3 r dcr3 r pcb3 + ?? i l3 ? = (eq. 8) v isen4 r dcr4 r pcb4 + ?? i l4 ? = (eq. 9)
isl95712 fn8566 rev 1.00 page 14 of 35 november 2, 2015 the isl95712 will adjust the phase pulse-width relative to the other phases to make v isen1 =v isen2 =v isen3 =v isen4 , thus to achieve i l1 =i l2 =i l3 =i l4 , when r dcr1 =r dcr2 =r dcr3 =r dcr4 and r pcb1 =r pcb2 =r pcb3 =r pcb4 . using the same components for l1, l2, l3 and l4 provides a good match of r dcr1 , r dcr2 , r dcr3 and r dcr4 . board layout determines r pcb1 , r pcb2 , r pcb3 and r pcb4 . it is recommended to have a symmetrical layout fo r the power delivery path between each inductor and the output voltage rail, such that r pcb1 =r pcb2 =r pcb3 =r pcb4 . sometimes, it is difficult to implement symmetrical layout. for the circuit shown in figure 13 , asymmetric layout causes different r pcb1 , r pcb2 , r pcb3 and r pcb4 values, thus creating a current imbalance. figure 14 shows a differential sensing current balancing circuit recommended for isl95712. the current sensing traces should be routed to the inductor pads so they only pick up the inductor dcr voltage. each isen pin sees the average voltage of three sources: its ow n, phase inductor phase-node pad, and the other two phase inductor output side pads. equations 10 through 13 give the isen pin voltages: the isl95712 will make v isen1 = v isen2 = v isen3 = v isen4 as shown in equations 14 and 16 : rewriting equation 14 gives equation 17 : rewriting equation 15 gives equation 18 : rewriting equation 16 gives equation 19 : combining equations 17 through 19 give: therefore: current balancing (i l1 =i l2 =i l3 =i l4 ) is achieved when r dcr1 =r dcr2 =r dcr3 =r dcr4 . r pcb1 , r pcb2 , r pcb3 and r pcb4 do not have any effect. since the slave ripple capacitor voltages mimic the inductor currents, the r3? modulator can naturally achieve excellent current balancing during steady state and dynamic operations. figure 15 shows the current balancing performance of a three-phase evaluation board with load transient of 12a/51a at different rep rates. the inductor currents follow the load current dynamic change with the output capacitors supplying the difference. the inductor currents can track the load current well at a low repetition rate, but canno t keep up when the repetition rate gets into the hundred-khz range, where it is out of the control loop bandwidth. the cont roller achieves excellent current balancing in all cases installed. figure 14. differential-sensing current balancing circuit internal to ic v o isen3 l3 r isen c isen isen2 r isen c isen isen1 r isen c isen l2 l1 r dcr3 r dcr2 r dcr1 phase3 phase2 phase1 i l3 i l2 i l1 r pcb3 r pcb2 r pcb1 r isen r isen r isen r isen r isen r isen v3p v 3n v2p v 2n v1p v 1n l4 r dcr4 i l4 r pcb4 v 4n v4p phase4 r isen c isen r isen r isen r isen r isen r isen r isen isen4 v isen1 v 1p v 2n v 3n v 4n +++ = (eq. 10) v isen2 v 1n v 2p v 3n v 4n +++ = (eq. 11) v isen3 v 1n v 2n v 3p v 4n +++ = (eq. 12) v isen4 v 1n v 2n v 3n v 4p +++ = (eq. 13) v 1p v 2n v 3n v + 4n ++ v 1n v 2p v 3n v 4n +++ = (eq. 14) v 1n v 2p v 3n v + 4n ++ v 1n v 2n v 3p v 4n +++ = (eq. 15) v 1n v 2n v 3p v + 4n ++ v 1n v 2n v 3n v 4p +++ = (eq. 16) v 1p v 1n C v 2p v 2n C = (eq. 17) v 2p v 2n C v 3p v 3n C = (eq. 18) v 3p v 3n C v 4p v 4n C = (eq. 19) v 1p v 1n C v 2p v 2n C v 3p v 3n C v 4p v 4n C === (eq. 20) r dcr1 i l1 ? r dcr2 i l2 ? r dcr3 i l3 ? r dcr4 i l4 ? === (eq. 21)
isl95712 fn8566 rev 1.00 page 15 of 35 november 2, 2015 modes of operation the core vr can be configured for 4-, 3-, 2- or 1-phase operation. table 1 shows core vr configuratio ns and operational modes, programmed by the isen4, isen3 and isen2 pin status and the psi0_l and psi1_l commands via the svi 2 interface. the svi 2 interface description of these bits is outlined in table 9 . the isenx pins disable the channel which they are related to. for example, to setup a 3-phase config uration the isen4 pin is tied to 5v. this disables channel 4 of the controller on the core side. in a 3-phase configuration, the core vr operates in 3-phase ccm, with psi0_l and psi_l both high. if psi0_l is taken low via the svi 2 interface, the core vr sheds phase 3. the core vr then operates 2-phase and remains in ccm. when both psi0_l and psi1_l are taken low, the core vr sheds phase 2 and the core vr enters 1-phase diode emulation (de) mode. for 2-phase configurations, the core vr operates in 2-phase ccm with psi0_l and psi_l both high. if psi0_l is taken low via the svi 2 interface, the core vr sheds phase 2 and the core vr operates in 1-phase and remains in ccm. when both psi0_l and psi1_l are taken low, the core vr operates in 1-phase de mode. in a 1-phase configuration, the core vr operates in 1-phase ccm and remains in this mode when ps i0_l is taken low. when both psi0_l and psi1_l are taken low, the controller enters de mode. when the core vr is taken into psi1 mode, where both psi0_l and psi1_l are taken low, the isl9 5712 will shed any additional phases in excess of phase 1. if there is a vid change as well, the regulator will then slew the output to the new vid level in ccm mode. once the output has reached the new vid level, the core vr is then placed into de mode . the core vr can be disabled completely by connecting isen1 to +5v. figure 15. current balancing during dynamic operation. ch1: i l1 , ch2: i load , ch3: i l2 , ch4: i l3 rep rate = 10khz rep rate = 25khz rep rate = 50khz rep rate = 100khz rep rate = 200khz
isl95712 fn8566 rev 1.00 page 16 of 35 november 2, 2015 the isl95712 northbridge vr can be configured for 3-, 2-, or 1- phase operation. table 2 shows the northbridge vr configurations and operational modes, which are programmed by the isen3_nb and isen2_nb pin status and the psi0_l and psi1_l bits of the svi 2 command. in a 1-phase configuration, the is en2_nb pin is tied to +5v. the northbridge vr operates in 1-phase ccm when both psi0_l and psi1_l are high and continues in this mode when psi0_l is taken low. the controller enters 1-phase de mode when both psi0_l and psi1_l are low. when the northbridge vr is taken into psi1 mode, where both psi0_l and psi1_l are taken low, the isl95712 will shed any additional phases in excess of ph ase 1. if there is a vid change as well, the regulator will then slew the output to the new vid level in ccm mode. once the output has reached the new vid level, the northbridge vr is then placed into de mode. the northbridge vr can be disabled completely by tying isen1_nb to 5v. dynamic operation core and northbridge vrs behave the same during dynamic operation. the controller responds to vid-on-the-fly changes by slewing to the new voltage at the slew rate programmed, see table 4 . during negative vid transitions, the output voltage decays to the lower vid value at the slew rate determined by the load. the r3? modulator intrinsically has voltage feed-forward. the output voltage is insensitive to a fast slew rate input voltage change. adaptive body diode conduction time reduction in dcm, the controller turns off the low-side mosfet when the inductor current approaches zero. during on-time of the low-side mosfet, phase voltage is negative and the amount is the mosfet r ds(on) voltage drop, which is proportional to the inductor current. a phase comparator inside the controller monitors the phase voltage during on-time of the low-side mosfet and compares it with a threshold to determine the zero crossing point of the inductor current. if the inductor current has not reached zero when the low-side mosfet turns off, it will flow through the low-side mosfet bo dy diode, causing the phase node to have a larger voltage drop until it decays to zero. if the inductor current has crossed zero and reversed the direction when the low-side mosfet turns off, it will flow through the high-side mosfet body diode, causing the phase node to have a spike until it decays to zero. th e controller continues monitoring the phase voltage after turning off the low-side mosfet. to minimize the body diode-related loss, the controller also adjusts the phase comparator threshold vo ltage accordingly in iterative steps such that the low-side mosfet body diode conducts for approximately 40ns. resistor configuration options the isl95712 uses the comp, comp_nb and prog pins to configure some function ality within the ic. resistors from these pins to gnd are read during the first portion of the soft-start sequence. the following sections outline how to select the resistor values for each of thes e pins to correctly program the output voltage offset of each output, vid-on-the-fly slew rate and switching frequency used for both vrs. vr offset programming a positive or negative offset is programmed for the core vr using a resistor to ground from the co mp pin and the northbridge in a similar manner from the comp_nb pin. table 3 provides the resistor value to select the desired output voltage offset. the 1% tolerance resistor value shown in table 3 must be used to program the corresponding core or nb ou tput voltage offset. the min and max tolerance values provide margin to insure the 1% tolerance resistor will be read correctly. table 2. northbridge vr modes of operation config. isen3 _nb isen2_nb psi0_l and psi1_l mode 3-phase nb vr configuration to power stage to power stage 11 2-phase ccm 01 1-phase ccm 00 1-phase de 2-phase nb vr configuration tied to 5v to power stage 11 2-phase ccm 01 1-phase ccm 00 1-phase de 1-phase nb vr configuration tied to 5v tied to 5v 11 1-phase ccm 01 1-phase ccm 00 1-phase de table 3. comp and comp_nb outp ut voltage offset selection resistor value [ k ] comp v core offset [mv] comp_nb offset [mv] min tolerance 1% tolerance value max tolerance 3.96 4.02 4.07 -43.75 18.75 7.76 7.87 7.98 -37.5 31 .25 11.33 11.5 11.67 -31.25 43.76 16.65 16.9 17.15 -25 50 19.3 19.6 19.89 -18.75 37.5 24.53 24.9 25.27 -12.5 25 33.49 34.0 34.51 -6.25 12.5 40.58 41.2 41.81 6.25 0 51.52 52.3 53.08 18.75 18.75 72.10 73.2 74.29 31.25 31.25 93.87 95.3 96.72 43.76 43.76 119.19 121 112.81 50 50 151.69 154 156.31 37.5 37.5 179.27 182 184.73 25 25 206.85 210 213.15 12.5 12.5 open 00
isl95712 fn8566 rev 1.00 page 17 of 35 november 2, 2015 vid-on-the-fly slew rate selection the prog resistor is used to select the slew rate for vid changes commanded by the processor. once selected, the slew rate is locked in during soft-start and is not adjustable during operation. the lowest slew rate that can be selected is 10mv/s, which is above the minimum of 7.5mv/s required by the svi2 specification. the slew rate selected sets the slew rate for both core and northbridge vrs. the controller does not allow for independent selection of slew rate. ccm switching frequency the core and northbridge vr switching frequency is set by the programming resistors on comp_nb and prog. when the isl95712 is in continuous conduction mode (ccm), the switching frequency is not absolutely constant due to the nature of the r3? modulator. as explained in ? multiphase r3? modulator ? on page 10 , the effective switching frequency increases during load insertio n and decreases during load release to achieve fast response. thus, the switching frequency is relatively constant at steady state. variation is expected when the power stage condition, such as input voltage, output voltage, load, etc. changes. the variation is usually less than 10% and does not have any significant effect on output voltage ripple magnitude. table 5 defines the switching frequency based on the resistor values used to prog ram the comp_nb and prog pins. use the previous tables related to comp_nb and prog to determine the correct resistor value in these ranges to program the desired output offset and slew rate. the controller monitors svi commands to determine when to enter power-saving mode, implement dynamic vid changes and shut down individual outputs. amd serial vid interface 2.0 the on-board serial vid interface 2.0 (svi 2) circuitry allows the amd processor to directly cont rol the core and northbridge voltage reference levels within the isl95712. once the pwrok signal goes high, the ic begins monitoring the svc and svd pins for instructions. the isl95712 uses a digital-to-analog converter (dac) to generate a reference voltage based on the decoded svi value. see figure 10 on page 12 for a simple svi interface timing diagram. pre-pwrok metal vid typical motherboard start-up begi ns with the controller decoding the svc and svd inputs to determine the pre-pwrok metal vid setting (see table 6 ). once the enable in put exceeds the rising threshold, the isl95712 decodes and locks the decoded value into an on-board hold register. once the programming pins are re ad, the internal dac circuitry begins to ramp core and northbridge vrs to the decoded pre-pwrok metal vid output level. the digital soft-start circuitry ramps the internal reference to the target gradually at a fixed rate of approximately 5mv/s un til the output voltage reaches ~250mv and then at the programm ed slew rate. the controlled ramp of all output voltage planes reduces inrush current during the soft-start interval. at the end of the soft-start interval, the pgood and pgood_nb ou tputs transition high, indicating both output planes are within regulation limits. if the enable input falls below th e enable falling threshold, the isl95712 tri-states both outputs. pgood and pgood_nb are pulled low with the loss of enable. the core and northbridge vr output voltages decay, based on output capacitance and load table 4. prog resistor selection resistor value [ k ] slew rate for core and northbridge [mv/s] 4.02 20 7.87 15 11.5 12.5 16.9 10 19.6 20 24.9 15 34.0 12.5 41.2 10 52.3 20 73.2 15 95.3 12.5 121 10 154 20 182 15 210 12.5 open 10 table 5. switching frequency selection frequency [khz] comp_nb range [k ] prog range [k ] 300 57.6 to open 19.1 to 41.2 or 154 to open 350 4.02 to 41.2 19.1 to 41.2 or 154 to open 400 57.6 to open 5.62 to 16.9 or 57.6 to 121 450 4.02 to 41.2 5.62 to 16.9 or 57.6 to 121 table 6. pre-pwrok metal vid codes svc svd output voltage (v) 00 1.1 01 1.0 1 0 0.9 1 1 0.8
isl95712 fn8566 rev 1.00 page 18 of 35 november 2, 2015 leakage resistance. if bias to vd d falls below the por level, the the isl95712 responds in the manner previously described. once vdd and enable rise above their respective rising thresholds, the internal dac circuitry reacquires a pre-pwrok metal vid code, and the controller soft-starts. svi interface active once the core and northbridge vrs have successfully soft-started and pgood and pgood_nb signal s transition high, pwrok can be asserted externally to the isl95712. once pwrok is asserted to the ic, svi instructions can begin as the controller actively monitors the svi interface. details of the svi bus protocol are provided in the ?amd serial vid interface 2.0 (svi2) specification?. see amd publication #48022. once a vid change command is received, the isl95712 decodes the information to determine which vr is affected and the vid target is determined by the byte combinations in table 7 . the internal dac circuitry steps the output voltage of the vr commanded to the new vid level. during this time, one or more of the vr outputs could be targeted. in the event either vr is commanded to power-off by serial vid commands, the pgood signal remains asserted. if the pwrok input is deasserted, then the controller steps both the core and the northbridge vrs back to the stored pre-pwrok metal vid level in the holding register from initial soft-start. no attempt is made to read the svc and svd inputs during this time. if pwrok is reasserted, then the isl95712 svi interface waits for instructions. if enable goes low during no rmal operation, all external mosfets are tri-stated and both pgood and pgood_nb are pulled low. this event clears the pre-pwrok metal vid code and forces the controller to check svc and svd upon restart, storing the pre-pwrok metal vid code found on restart. a por event on vcc during normal operation shuts down both regulators, and both pgood outputs are pulled low. the pre-pwrok metal vid code is not retained. loss of vin during operation will typically cause the controller to enter a fault condition on one or both outputs as the output voltage collapses. the controller will shut down bo th core and northbridge vrs and latch off. the pre-pwrok metal vid code is not retained during the process of cycling enable to reset the fault latch and restart the controller. vid-on-the-fly transition once pwrok is high, the isl95712 detects this flag and begins monitoring the svc and svd pins for svi instructions. the microprocessor follows the protoc ol outlined in the following sections to send instructions for vid-on-the-fly transitions. the isl95712 decodes the instruction and acknowledges the new vid code. for vid codes higher than the current vid level, the isl95712 begins stepping the commanded vr outputs to the new vid target at the fixed slew rate of 10mv/s. once the dac ramps to the new vid code, a vid-on-the-fly complete (votfc) request is sent on the svi lines. when the vid codes are lower than the current vid level, the isl95712 checks the state of power state bits in the svi command. if power state bits are not active, the controller begins stepping the regulator output to the new vid target. if the power state bits are active, the controller allows the output voltage to decay and slowly steps the dac down with the natural decay of the output. this allows the controller to quickly recover and move to a high vid code if commanded. the controller issues a votfc request on the svi lines once the svi command is decoded and prior to reaching the final output voltage. votfc requests do not take priority over telemetry per the amd svi 2 specification. svi data communication protocol the svi wire protocol is based on the i 2 c bus concept. two wires [serial clock (svc) and serial data (svd)], carry information between the amd processor (master) and vr controller (slave) on the bus. the master initiates and terminates svi transactions and drives the clock, svc, during a transaction. the amd processor is always the master and the voltage regulators are the slaves. the slave receives the svi transactions and acts accordingly. mobile svi wire protocol timing is based on high-speed mode i 2 c. see amd publication #48022 for additional details.
isl95712 fn8566 rev 1.00 page 19 of 35 november 2, 2015 table 7. serial vid codes svid[7:0] voltage (v) svid[6:0] voltage (v) svid[6:0] voltage (v) svid[6:0] voltage (v) 0000_0000 1.55000 0010_0000 1.35000 0100_0000 1.15000 0110_0000 0.95000 0000_0001 1.54375 0010_0001 1.34375 0100_0001 1.14375 0110_0001 0.94375 0000_0010 1.53750 0010_0010 1.33750 0100_0010 1.13750 0110_0010 0.93750 0000_0011 1.53125 0010_0011 1.33125 0100_0011 1.13125 0110_0011 0.93125 0000_0100 1.52500 0010_0100 1.32500 0100_0100 1.12500 0110_0100 0.92500 0000_0101 1.51875 0010_0101 1.31875 0100_0101 1.11875 0110_0101 0.91875 0000_0110 1.51250 0010_0110 1.31250 0100_0110 1.11250 0110_0110 0.91250 0000_0111 1.50625 0010_0111 1.30625 0100_0111 1.10625 0110_0111 0.90625 0000_1000 1.50000 0010_1000 1.30000 0100_1000 1.10000 0110_1000 0.90000 0000_1001 1.49375 0010_1001 1.29375 0100_1001 1.09375 0110_1001 0.89375 0000_1010 1.48750 0010_1010 1.28750 0100_1010 1.08750 0110_1010 0.88750 0000_1011 1.48125 0010_1011 1.28125 0100_1011 1.08125 0110_1011 0.88125 0000_1100 1.47500 0010_1100 1.27500 0100_1100 1.07500 0110_1100 0.87500 0000_1101 1.46875 0010_1101 1.26875 0100_1101 1.06875 0110_1101 0.86875 0000_1110 1.46250 0010_1110 1.26250 0100_1110 1.06250 0110_1110 0.86250 0000_1111 1.45625 0010_1111 1.25625 0100_1111 1.05625 0110_1111 0.85625 0001_0000 1.45000 0011_0000 1.25000 0101_0000 1.05000 0111_0000 0.85000 0001_0001 1.44375 0011_0001 1.24375 0101_0001 1.04375 0111_0001 0.84375 0001_0010 1.43750 0011_0010 1.23750 0101_0010 1.03750 0111_0010 0.83750 0001_0011 1.43125 0011_0011 1.23125 0101_0011 1.03125 0111_0011 0.83125 0001_0100 1.42500 0011_0100 1.22500 0101_0100 1.02500 0111_0100 0.82500 0001_0101 1.41875 0011_0101 1.21875 0101_0101 1.01875 0111_0101 0.81875 0001_0110 1.41250 0011_0110 1.21250 0101_0110 1.01250 0111_0110 0.81250 0001_0111 1.40625 0011_0111 1.20625 0101_0111 1.00625 0111_0111 0.80625 0001_1000 1.40000 0011_1000 1.20000 0101_1000 1.00000 0111_1000 0.80000 0001_1001 1.39375 0011_1001 1.19375 0101_1001 0.99375 0111_1001 0.79375 0001_1010 1.38750 0011_1010 1.18750 0101_1010 0.98750 0111_1010 0.78750 0001_1011 1.38125 0011_1011 1.18125 0101_1011 0.98125 0111_1011 0.78125 0001_1100 1.37500 0011_1100 1.17500 0101_1100 0.97500 0111_1100 0.77500 0001_1101 1.36875 0011_1101 1.16875 0101_1101 0.96875 0111_1101 0.76875 0001_1110 1.36250 0011_1110 1.16250 0101_1110 0.96250 0111_1110 0.76250 0001_1111 1.35625 0011_1111 1.15625 0101_1111 0.95625 0111_1111 0.75625 1000_0000 0.75000 1010_0000 0.55000* 1100_0000 0.35000* 1110_0000 0.15000* 1000_0001 0.74375 1010_0001 0.54375* 1100_0001 0.34375* 1110_0001 0.14375* 1000_0010 0.73750 1010_0010 0.53750* 1100_0010 0.33750* 1110_0010 0.13750* 1000_0011 0.73125 1010_0011 0.53125* 1100_0011 0.33125* 1110_0011 0.13125* 1000_0100 0.72500 1010_0100 0.52500* 1100_0100 0.32500* 1110_0100 0.12500* 1000_0101 0.71875 1010_0101 0.51875* 1100_0101 0.31875* 1110_0101 0.11875* 1000_0110 0.71250 1010_0110 0.51250* 1100_0110 0.31250* 1110_0110 0.11250* 1000_0111 0.70625 1010_0111 0.50625* 1100_0111 0.30625* 1110_0111 0.10625*
isl95712 fn8566 rev 1.00 page 20 of 35 november 2, 2015 1000_1000 0.70000 1010_1000 0.50000* 1100_1000 0.30000* 1110_1000 0.10000* 1000_1001 0.69375 1010_1001 0.49375* 1100_1001 0.29375* 1110_1001 0.09375* 1000_1010 0.68750 1010_1010 0.48750* 1100_1010 0.28750* 1110_1010 0.08750* 1000_1011 0.68125 1010_1011 0.48125* 1100_1011 0.28125* 1110_1011 0.08125* 1000_1100 0.67500 1010_1100 0.47500* 1100_1100 0.27500* 1110_1100 0.07500* 1000_1101 0.66875 1010_1101 0.46875* 1100_1101 0.26875* 1110_1101 0.06875* 1000_1110 0.66250 1010_1110 0.46250* 1100_1110 0.26250* 1110_1110 0.06250* 1000_1111 0.65625 1010_1111 0.45625* 1100_1111 0.25625* 1110_1111 0.05625* 1001_0000 0.65000 1011_0000 0.45000* 1101_0000 0.25000* 1111_0000 0.05000* 1001_0001 0.64375 1011_0001 0.44375* 1101_0001 0.24375* 1111_0001 0.04375* 1001_0010 0.63750 1011_0010 0.43750* 1101_0010 0.23750* 1111_0010 0.03750* 1001_0011 0.63125 1011_0011 0.43125* 1101_0011 0.23125* 1111_0011 0.03125* 1001_0100 0.62500 1011_0100 0.42500* 1101_0100 0.22500* 1111_0100 0.02500* 1001_0101 0.61875 1011_0101 0.41875* 1101_0101 0.21875* 1111_0101 0.01875* 1001_0110 0.61250 1011_0110 0.41250* 1101_0110 0.21250* 1111_0110 0.01250* 1001_0111 0.60625 1011_0111 0.40625* 1101_0111* 0.20625* 1111_0111 0.00625* 1001_1000 0.60000* 1011_1000 0.40000* 1101_1000 0.20000* 1111_1000 off* 1001_1001 0.59375* 1011_1001 0.39375* 1101_1001 0.19375* 1111_1001 off* 1001_1010 0.58750* 1011_1010 0.38750* 1101_1010 0.18750* 1111_1010 off* 1001_1011 0.58125* 1011_1011 0.38125* 1101_1011 0.18125* 1111_1011 off* 1001_1100 0.57500* 1011_1100 0.37500* 1101_1100 0.17500* 1111_1100 off* 1001_1101 0.56875* 1011_1101 0.36875* 1101_1101 0.16875* 1111_1101 off* 1001_1110 0.56250* 1011_1110 0.36250* 1101_1110 0.16250* 1111_1110 off* 1001_1111 0.55625* 1011_1111 0.35625* 1101_1111 0.15625* 1111_1111 off* note: * indicates a vid not required for amd family 10h proc essors. loosened amd requir ements at these levels. table 7. serial vid codes (continued) svid[7:0] voltage (v) svid[6:0] voltage (v) svid[6:0] voltage (v) svid[6:0] voltage (v)
isl95712 fn8566 rev 1.00 page 21 of 35 november 2, 2015 svi bus protocol the amd processor bus protocol is similar to smbus send byte protocol for vid transactions. the amd svd packet structure is shown in figure 16 . the description of each bit of the three bytes that make up the svi command are shown in table 8 . during a transaction, the processor sends the start sequence followed by each of the three bytes, which end with an optional acknowledge bit. the isl95712 does not drive the svd line during the ack bit. finally, the processor sends the stop sequence. after the isl95712 has detected the stop, it can then proceed with the commanded action from the transaction. power states svi2 defines two power state indicator levels, see tables 1 , 2 , and 9 . as processor current consumption is reduced, the power state indicator level changes to improve vr efficiency under low power conditions. for the core vr operating in 4-phase mode (when psi0_l is asserted) channels 3 and 4 ar e tri-stated. the controller continues to operate in 2-phase ccm. the shedding of phases improves the efficiency of the vr at the light to moderate load levels of the cpu in this power state. when psi1_l is asserted the core vr sheds channel 2. if there is a corresponding vid change, then the output is moved to the new vid level while in single phase de mode. once the output is at the proper vid level, channel 1 enters diode emulation mode to further boost light-load efficiency in this power state. for the northbridge vr operating in 3-phase mode, when psi0_l is asserted, channels 2 and 3 are tri-stated while channel 1 continues in continuous conduc tion mode. when psi1_l is asserted, the output is moved to the new vid level if one is commanded and channel 1 then enters diode emulation mode to conserve power. it is possible for the processor to assert or deassert psi0_l and psi1_l out of order. psi0_l takes priority over psi1_l. if psi0_l is deasserted while psi1_l is still asserted, the isl95712 will return the selected vr back full channel ccm operation. for example, if the core vr is conf igured for 4-phase operation and both psi0_l and psi1_l are asse rted low during a command, the vr will shed three phases and op erate in 1-phase de mode. if an svi command follows which takes psi0_l high, but leaves psi1_l low, the vr will exit power savings mode and being operation in 4-phase ccm mode. figure 16. svd packet structure 1 2 3 4 5 6 7 12 14151617 13 10 svd svc s t a r t p s i 1 _ l vid bits [7:1] 11 89 18 19 20 21 22 23 24 25 26 27 vid bit [0] p s i 0 _ l ack ack ack
isl95712 fn8566 rev 1.00 page 22 of 35 november 2, 2015 dynamic load line slope trim the isl95712 supports the svi2 ability for the processor to manipulate the load line slope of the core and northbridge vrs independently using the serial vid interface. the slope manipulation applies to the initial load line slope. a load line slope trim will typically coincide with a votf change. see table 10 for more information about the load line slope trim feature of the isl95712. the disable ll selection is not recommended unless operation without a ll is required and considered during the co mpensation of the vr. dynamic offset trim the isl95712 supports the svi2 ability for the processor to manipulate the output voltage offset of the core and northbridge vrs. this offset is in addition to any output voltage offset set via the comp resistor reader. the dy namic offset trim can disable the comp resistor programmed of fset of either output when disable all offset is selected. telemetry the isl95712 can provide voltage and current information to the amd cpu through the telemetry system outlined by the amd svi2 specification. the telemetry data is transmitted through the svc and svt lines of the svi2 interface. current telemetry is based on a voltage generated across a 133k resistor placed from the imon pin to gnd. the current flowing out of the imon pin is proportional to the load current in the vr. the i sum current defined in ? voltage regulation and load line implementation ? on page 12 , provides the base conversion from the load current to the internal amplifier created i sum current. the i sum current is then divided down by a factor of 4 to create the imon current, which flows out of the imon pin. the i sum current will measure 36a when the load current is at full load based on a droop current designed for 45a at the same load current. the difference between the i sum current and the droop current is provided in equation 2 . the imon current will measure 11.25a at full load cu rrent for the vr and the imon voltage will be 1.2v. the load percentage, which is reported by the ic is based on the this voltage. when the load is 25% of the full load, the voltage on the imon pin will be 25% of 1.2v or 0.3v. the svi interface allows the selection of no telemetry, voltage only, or voltage and current telemetry on either or both of the vr outputs. the tfn bit along with the core and northbridge domain selector bits are used by the processor to change the functionality of telemetry, see table 12 for more information. table 10. load line slope trim definition load line slope trim [2:0] description 000 disable ll 001 -40% m change 010 -20% m change 011 no change 100 +20% m change 101 +40% m change 110 +60% m change 111 +80% m change table 11. offset trim definition offset trim [1:0] description 00 disable all offset 01 -25mv change 10 0mv change 11 +25mv change table 12. tfn truth table tfn, core, nb bits [21, 6, 7] description 1,0,1 telemetry is in voltage and current mode. therefore, voltage and current are sent for vdd and vddnb domains by the controller. 1,0,0 telemetry is in voltage mode only. only the voltage of vdd and vddnb domains is sent by the controller. 1,1,0 telemetry is disabled. 1,1,1 reserved
isl95712 fn8566 rev 1.00 page 23 of 35 november 2, 2015 pmbus interface the isl95712 includes a pmbus interface, which allows for user programmability of numerous operating parameters and for monitoring various parameters of the core and nb regulators. the pmbus address for the isl95712 is 1001111. table 13. pmbus read and write registers command code access default command name description 9bh r 01h manufacturer revision silicon revision starts at 01h d0h reserved d1h reserved d2h r/w 00h fault_status_2 bit bit value 01 5 (read only) isl95712 enabled isl95712 fault disabled 4 no fault core ov 3nb ov 2core ocp 1nb ocp 0 cml. indicates that an unsupported command is received or a write command to a read-only register or pec does not match d3h r xxh read_vout_core read the core voltage in adc format. each lsb is 6.25mv d4h r xxh read_iout_core read core current in adc format. ffh = 100% (7.5a on imon) d5h reserved d6h r xxh read_vout_nb read the nb voltag e in adc format. each lsb is 6.25mv d7h r xxh read_iout_nb read nb load current in adc format. ffh = 100% (7.5a on imon) d8h reserved d9h reserved dah reserved dbh reserved dch reserved ddh reserved deh r/w 00h lock_svid bit[0] value functionality 0 execute svi2 commands. pmbus commands dfh through e4h are not executed. these registers can still be read and written to. 1 execute pmbus commands dfh through e4h while ignoring svi2 commands. dfh r/w 08h set_vid_core set core vid, default se t to 800mv. each lsb is 6.25mv. metal vid level is determined by svc/svd logic levels at power-up. e0h r/w 00h offset_core set core offset. the offset range is from -250mv to +200mv. this is a 2?s complement number. bit[7] is the sign bit.
isl95712 fn8566 rev 1.00 page 24 of 35 november 2, 2015 protection features core vr and northbridge vr both provide overcurrent, current-balance, underv oltage and overvoltage fault protections. the controller also provides ov er-temperature protection. the following discussion is based on core vr and also applies to the northbridge vr. overcurrent the imon voltage provides a means of determining the load current at any moment in time. the overcurrent protection (ocp) circuitry monitors the imon voltage to determine when a fault occurs. based on the pr evious description in ? voltage regulation and load line implementation ? on page 12 , the current which flows out of the imon pin is proportional to the i sum current. the i sum current is created from the sensed voltage across c n , which is a measure of the load current based upon the sensing element selected. the imon current is gene rated internally and is 1/4 of the i sum current. the edc or iddspi ke current value for the amd cpu load is used to set the maximum current level for droop and the imon voltage of 1.2v, which indicates 100% loading for telemetry. the i sum current level at maximu m load, or iddspike, is 36a and this translates to an imon current level of 9a. the imon resistor is 133k and the 9a flowing through the imon resistor results in a 1.2v level at maximum loading of the vr. the overcurrent threshold is 1.5v on the imon pin. based on a 1.2v imon voltage equating to 10 0% loading, the additional 0.3v provided above this level equates to a 25% increase in load current before an ocp fault is detected. the edc or iddspike current is used to set the 1.2v on imon fo r full load current. thus the ocp level is 1.25 times the edc or iddspike current level. this additional margin above the edc or iddspike current allows the amd cpu to enter and exit the iddspike performance mode without issue unless the load current is out of line with the iddspike expectation, thus the need for overcurrent protection. when the voltage on the imon pin meets the overcurrent threshold of 1.5v, this triggers an ocp event. within 2s of detecting an ocp event, the cont roller asserts vr_hot_l low to communicate to the amd cpu to throttle back. a fault timer begins counting while imon is at or above the 1.5v threshold. the fault timer lasts 7.5s to 11s and then the controller takes action by tri-stating the active channels . this provides the cpu time to recover and reduce the load current. if the ocp conditions are relieved, then the fault timer is cleared and vr_hot_l is taken high clearing the fault condition. if the load current is not reduced and the ocp condition is maintain ed, the output voltage will fall below the undervoltage threshold du e to the lack of switching or a way-overcurrent fault could occur. either of these fault conditions will cause the controller to drop pgood of that output. when pgood is taken low, a fault flag from this vr is sent to the other vr and it is shut down within 1 0s and pgood of the other output is taken low. the isl95712 also features a way-overcurrent [woc] feature, which immediately takes the controller into shutdown. this protection is also referred to as fast overcurrent protection for short-circuit protection. if the imon current reaches 15a, woc is triggered. active channels are tri-stated and the controller is placed in shutdown and pgood is pulled low. there is no fault timer on the woc fault, the contro ller takes immediate action. the other controller output is also shut down within 10s. current-balance the controller monitors the isenx pin voltages to determine current-balance protection. if the isenx pin voltage difference is greater than 9mv for 1ms, the controller will declare a fault and latch off. e1h r/w 0fh loadline_pwrstate_core bit functionality [4:2] load line slope trim. refer to table 10 for proper usage. 1 sets psi0 power state. refer to table 9 for proper usage. 0 sets psi1 power state. refer to table 9 for proper usage. e2h r/w 08h set_vid_nb set nb vid, default set to 800mv. each lsb is 6.25mv. metal vid level is determined by svc/svd logic levels at power-up. e3h r/w 00h offset_nb set nb offset. the offset range is from -250mv to +200mv. this is a 2?s complement number. bit[7] is the sign bit. e4h r/w 0fh loadline_pwrstate_nb bit functionality [4:2] load line slope trim. refer to table 10 for proper usage. 1 sets psi0 power state. refer to table 9 for proper usage. 0 sets psi1 power state. refer to table 9 for proper usage. table 13. pmbus read and write registers (continued) command code access default command name description
isl95712 fn8566 rev 1.00 page 25 of 35 november 2, 2015 undervoltage if the vsen voltage falls below the output voltage vid value plus any programmed offsets by -325m v, the controller declares an undervoltage fault. the controller deasserts pgood and tri-states the power mosfets. overvoltage if the vsen voltage exceeds the output voltage vid value plus any programmed offsets by +325mv, the controller declares an overvoltage fault. the controller deasserts pgood and turns on the low-side power mosfets. the low-side power mosfets remain on until the output voltage is pulled down below the vid set value. once the output voltag e is below this level, the lower gate is tri-stated. if the output voltage rises above the overvoltage threshold again, the protection process is repeated. this behavior provides the maximum amount of protection against shorted high-side power mosfets while preventing output ringing below ground. thermal monitor [ntc, ntc_nb] the isl95712 features two thermal monitors using an external resistor network, which includes an ntc thermistor to monitor motherboard temperature and al ert the amd cpu of a thermal issue. figure 17 shows the basic thermal monitor circuit on the core vr ntc pin. the northbridg e vr features the same thermal monitor. the controller drives a 30a current out of the ntc pin and monitors the voltage at the pin. the current flowing out of the ntc pin creates a voltage th at is compared to a warning threshold of 640mv. when the voltage at the ntc pin falls to this warning threshold or below, the controller asserts vr_hot_l to alert the amd cpu to throttle back load current to stabilize the motherboard temperature. a thermal fault counter begins counting toward a minimum sh utdown time of 100s. the thermal fault counter is an up/down counter, so if the voltage at the ntc pin rises above the warning threshold, it will count down and extend the time for a thermal fault to occur. the warning threshold does have 20mv of hysteresis. if the voltage at the ntc pin continues to fall down to the shutdown threshold of 580mv or below, the controller goes into shutdown and triggers a thermal fault. the pgood pin is pulled low and tri-states the power mosfet s. a fault on either side will shutdown both vrs. as the board temperature rises, the ntc thermistor resistance decreases and the voltage at the ntc pin drops. when the voltage on the ntc pin drops below the over-temperature trip threshold, then vr_hot is pulled low. the vr_hot signal is used to change the cpu operation an d decrease power consumption. with the reduction in power consumption by the cpu, the board temperature decreases and the nt c thermistor voltage rises. once the over-temperature threshold is tripped and vr_hot is taken low, the over-temperature threshold changes to the reset level. the addition of hysteresis to the over-temperature threshold prevents nuisance trips. once both pin voltages exceed the over-temperature reset thresh old, the pull-down on vr_hot is released. the signal changes state and the cpu resumes normal operation. the over-tempe rature threshold returns to the trip level. table 14 summarizes the fault protections. table 14. fault protection summary fault type fault duration before protection protection action fault reset overcurrent 7.5s to 11.5s pwm tri-state enable toggle or vdd toggle phase current unbalance 1ms pwm tri-state, pgood latched low way-overcurrent (1.5xoc) immediately undervoltage -325mv pgood latched low. pwm tri-state. overvoltage +325mv pgood latched low. actively pulls the output voltage to below vid value, then tri-state. ntc thermal 100s min pgood latched low. pwm tri-state. ntc r ntc v ntc - + 30a internal to isl95712 figure 17. circuitry associated with the thermal monitor feature of the isl95712 r s monitor r +v r p warning 640mv shutdown 580mv vr_hot_l
isl95712 fn8566 rev 1.00 page 26 of 35 november 2, 2015 fault recovery all of the previously described fault conditions can be reset by bringing enable low or by bringing vdd below the por threshold. when enable and vdd return to their high operating levels, the controller resets the faults and soft-start occurs. interface pin protection the svc and svd pins feature protection diodes, which must be considered when removing power to vdd and vddio, but leaving it applied to these pins. figure 18 shows the basic protection on the pins. if svc and/or svd are powered but vdd is not, leakage current will flow from these pins to vdd. key component selection inductor dcr current-sensing network figure 19 shows the inductor dcr current-sensing network for a 4-phase solution. an inductor current flows through the dcr and creates a voltage drop. each inductor has two resistors in r sum and r o connected to the pads to accurately sense the inductor current by sensing the dcr voltage drop. the r sum and r o resistors are connected in a summing network as shown and feed the total current information to the ntc network (consisting of r ntcs , r ntc and r p ) and capacitor c n . r ntc is a negative temperature coefficient (ntc) thermistor, used to temperature compensate the inductor dcr change. the inductor output side pads are electrically shorted in the schematic but have some parasitic impedance in actual board layout, which is why one cannot simply short them together for the current-sensing summing network. it is recommended to use 1 ~10 ? r o to create quality signals. since r o value is much smaller than the rest of the curr ent sensing circuit, the following analysis ignores it. the summed inductor current information is presented to the capacitor c n . equations 22 through 26 describe the frequency domain relationship between inductor total current i o (s) and c n voltage v cn (s): where n is the number of phases. transfer function a cs (s) always has unity gain at dc. the inductor dcr value increases as the wi nding temperature increases, giving higher reading of the inductor dc current. the ntc r ntc value decrease as its temperature decreases. proper selection of r sum , r ntcs , r p and r ntc parameters ensures that v cn represents the inductor total dc current over the temperature range of interest. there are many sets of parameters that can properly temperature-compensate the dcr change. since the ntc network and the r sum resistors form a voltage divider, v cn is always a fraction of the inductor dcr voltage. it is recommended to have a higher ratio of v cn to the inductor dcr voltage so the droop circuit has a higher signal level to work with. a typical set of parameters that provide good temperature compensation are: r sum = 3.65k , r p = 11k , r ntcs = 2.61k and r ntc = 10k (ert-j1vr103j). the ntc network parameters may need to be fine tuned on ac tual boards. one can apply full svc, svd internal to isl95712 figure 18. protection devices on the svc and svd pins gnd vdd cn r sum r o r ntcs r ntc r p dcr l dcr l r sum r o phase2 phase3 i o l phase1 r o r sum ri vcn + - l phase4 r sum r o dcr dcr i sum+ i sum- figure 19. dcr current-sensing network v cn s ?? r ntcnet r ntcnet r sum n -------------- - + ------------------------------------------ dcr n ------------- ? ?? ?? ?? ?? ?? i o s ?? ? a cs ? s ?? = (eq. 22) r ntcnet r ntcs r ntc + ?? r p ? r ntcs r ntc r p ++ ---------------------------------------------------- = (eq. 23) a cs s ?? 1 s ? l ------ - + 1 s ? sns ------------ - + ---------------------- - = (eq. 24) ? l dcr l ------------- = (eq. 25) ? sns 1 r ntcnet r sum n -------------- - ? r ntcnet r sum n -------------- - + ------------------------------------------ c n ? -------------------------------------------------------- = (eq. 26)
isl95712 fn8566 rev 1.00 page 27 of 35 november 2, 2015 load dc current and record the output voltage reading immediately; then record the ou tput voltage reading again when the board has reached the thermal steady state. a good ntc network can limit the output voltage drift to within 2mv. it is recommended to follow the intersil evaluation board layout and current sensing network parameters to minimize engineering time. v cn (s) also needs to represent real-time i o (s) for the controller to achieve good transient resp onse. transfer function a cs (s) has a pole ? sns and a zero w l . one needs to match ? l and ? sns so a cs (s) is unity gain at all frequencies. by forcing ? l equal to ? sns and solving for the solution, equation 27 gives c n value. for example, given n = 4, r sum = 3.65k , r p = 11k , r ntcs =2.61k , r ntc = 10k , dcr = 0.88m and l = 0.36h, equation 27 gives c n =0.518f. assuming the compensator design is correct, figure 20 shows the expected load transient response waveforms if c n is correctly selected. when the load current i core has a square change, the output voltage v core also has a square response. if c n value is too large or too small, v cn (s) does not accurately represent real-time i o (s) and worsens the transient response. figure 21 shows the load transient response when c n is too small. v core sags excessively upon load insertion and may create a system failure. figure 22 shows the transient response when c n is too large. v core is sluggish in drooping to its final value. there is excessive overshoot if load insertion occurs during this time, which may negatively affect the cpu reliability. figure 23 shows the output voltage ring-back problem during load transient response. the load current i o has a fast step change, but the inductor current i l cannot accurately follow. instead, i l responds in first-order system fa shion due to the nature of the current loop. the esr and esl effe ct of the output capacitors makes the output voltage v o dip quickly upon load current change. however, the controller regulates v o according to the droop current i droop , which is a real-time representation of i l ; therefore, it pulls v o back to the level dictated by i l , causing the ring-back problem. this phenomenon is not observed when the output capacitor has very low esr and esl, as is the case with all ceramic capacitors. figure 24 shows two optional circuits for reduction of the ring-back. c n is the capacitor used to match the inductor time constant. it usually takes the parallel of two (or more) capacitors to get the desired value. figure 24 shows that two capacitors (c n.1 and c n.2 ) are in parallel. resistor r n is an optional component to reduce the v o ring-back. at steady state, c n.1 + c n.2 provides the desired c n capacitance. at the beginning of i o c n l r ntcnet r sum n -------------- - ? r ntcnet r sum n -------------- - + ------------------------------------------ dcr ? -------------------------------------------------------------- - = (eq. 27) figure 20. desired load tr ansient response waveforms o i v o figure 21. load transient response when c n is too small o i v o figure 22. load transient response when c n is too large o i v o figure 23. output voltage ring-back problem o i v o l i ring back figure 24. optional circuits for ring-back reduction c n.2 r ntcs r ntc r p r i isum+ isum- r ip c ip optional v cn c n.1 r n optional + -
isl95712 fn8566 rev 1.00 page 28 of 35 november 2, 2015 change, the effective capacitance is less because r n increases the impedance of the c n.1 branch. as figure 21 shows, v o tends to dip when c n is too small, and this effect reduces the v o ring-back. this effect is more pronounced when c n.1 is much larger than c n.2 . it is also more pronounced when r n is bigger. however, the presence of r n increases the ripple of the v n signal if c n.2 is too small. it is recommended to keep c n.2 greater than 2200pf. r n value usually is a few ohms. c n.1 , c n.2 and r n values should be determined through tuni ng the load transient response waveforms on an actual board. r ip and c ip form an r-c branch in parallel with r i , providing a lower impedance path than r i at the beginning of i o change. r ip and c ip do not have any effect at steady state. through proper selection of r ip and c ip values, i droop can resemble i o rather than i l , and v o will not ring back. the recommended value for r ip is 100 . c ip should be determined through tuning the load transient response waveforms on an actual board. the recommended range for c ip is 100pf~2000pf. however, it should be noted that the r ip - c ip branch may distort the i droop waveform. instead of being triangular as the real inductor current, i droop may have sharp spikes, which may adversely affect i droop average value detection and therefore may affect ocp accuracy. user discretion is advised. resistor current-sensing network figure 25 shows the resistor current-sensing network for a 4-phase solution. each inductor has a series current sensing resistor, r sen . r sum and r o are connected to the r sen pads to accurately capture the inductor current information. the r sum and r o resistors are connected to capacitor c n . r sum and c n form a filter for noise attenuation. equations 28 through 30 give the v cn (s) expression. transfer function a rsen (s) always has unity gain at dc. current-sensing resistor r sen value does not have significant variation over-temperature, so there is no need for the ntc network. the recommended values are r sum = 1k and c n = 5600pf. overcurrent protection refer to equation 2 on page 13 and figures 19 and 25 ; resistor r i sets the i sum current, which is proportional to droop current and imon current. the ocp threshol d is 1.5v on the imon pin, which equates to an imon current of 11.25a using a 133k imon resistor. the corresponding i sum is 45a, which results in an i droop of 56.25a. at full load current, i omax , the i sum current is 36a and the resulting i droop is 45a. the ratio of i sum at ocp relative to full load current is 1.25. therefore, the ocp current trip level is 25% higher than the full load current. for inductor dcr sensing, equation 31 gives the dc relationship of v cn (s) and i o (s): substitution of equation 31 into equation 2 gives equation 32 : therefore: substitution of equation 23 and application of the ocp condition in equation 33 gives equation 34 : where i omax is the full load current and i droopmax is the corresponding droop current. for example, given n = 4, r sum = 3.65k , r p = 11k , r ntcs = 2.61k , r ntc = 10k , dcr = 0.88m , i omax = 100a and i droopmax = 45 a. equation 34 gives r i =529 . figure 25. resistor current-sensing network cn r sum r o dcr l dcr l r sum r o phase2 phase3 i o phase1 r o r sum r i vcn rsen rsen + - phase4 rsen rsen dcr l dcr l r sum r o i sum+ i sum- v cn s ?? r sen n ------------- i o s ?? ? a rsen ? s ?? = (eq. 28) a rsen s ?? 1 1 s ? sns ------------ - + ---------------------- - = (eq. 29) ? rsen 1 r sum n -------------- - c n ? ---------------------------- - = (eq. 30) v cn r ntcnet r ntcnet r sum n -------------- - + ------------------------------------------ dcr n ------------- ? ?? ?? ?? ?? ?? i o ? = (eq. 31) i droop 5 4 -- - 1 r i ----- r ntcnet r ntcnet r sum n -------------- - + ------------------------------------------ dcr n ------------- ? ? i o ? ? = (eq. 32) r i 5 4 -- - r ntcnet dcr ? i o ? nr ntcnet r sum n -------------- - + ?? ?? ? i droop ? --------------------------------------------------------------- ------------------ - ? = (eq. 33) r i 5 4 -- - r ntcs r ntc + ?? r p ? r ntcs r ntc r p ++ ---------------------------------------------------- dcr ? i omax ? n r ntcs r ntc + ?? r p ? r ntcs r ntc r p ++ ---------------------------------------------------- r sum n -------------- - + ?? ?? ?? ? i droopmax ? --------------------------------------------------------------- ------------------------------------------------------------- - ? = (eq. 34)
isl95712 fn8566 rev 1.00 page 29 of 35 november 2, 2015 for resistor sensing, equation 35 gives the dc relationship of v cn (s) and i o (s). substitution of equation 35 into equation 2 gives equation 36 : therefore: substitution of equation 37 and application of the ocp condition in equation 33 gives equation 38 : where i omax is the full load current and i droopmax is the corresponding droop current. for example, given n = 4, r sen =1m , i omax = 100a and i droopmax = 45a, equation 38 gives r i = 694 . load line slope see figure 12 for load line implementation. for inductor dcr sensing, substitution of equation 32 into equation 3 gives the load line slope expression: for resistor sensing, substitution of equation 36 into equation 3 gives the load line slope expression : substitution of equation 33 and rewriting equation 39 , or substitution of equation 37 and rewriting equation 40 , gives the same result as in equation 41 : one can use the full-load condition to calculate r droop . for example, given i omax = 100a, i droopmax = 45a and ll = 2.1m , equation 41 gives r droop = 4.67k . it is recommended to start with the r droop value calculated by equation 41 and fine-tune it on the actual board to get accurate load line slope. one should record the output volt age readings at no load and at full load for load line slope calculation. reading the output voltage at lighter load instead of full load will increase the measurement error. compensator figure 20 shows the desired load transient response waveforms. figure 26 shows the equivalent circuit of a voltage regulator (vr) with the droop function. a vr is equivalent to a voltage source (= vid) and output impedance z out (s). if z out (s) is equal to the load line slope ll, i.e., a cons tant output impedance, then in the entire frequency range, v o will have a square response when i o has a square change. intersil provides a microsoft excel-based spreadsheet to help design the compensator and the cu rrent sensing network so that vr achieves constant output impedance as a stable system. a vr with active droop function is a dual-loop system consisting of a voltage loop and a droop loop, which is a current loop. however, neither loop alone is sufficient to describe the entire system. the spreadsheet shows two loop gain transfer functions, t1(s) and t2(s), that describe the entire system. figure 27 conceptually shows t1(s) measurement set-up, and figure 28 conceptually shows t2(s) measurement set-up. the vr senses the inductor current, multiplies it by a gain of th e load line slope, adds it on top of the sensed output voltage, and then feeds it to the compensator. t1 is measured af ter the summing node, and t2 is measured in the voltage loop before the summing node. the spreadsheet gives both t1(s) and t2(s) plots. however, only t2(s) can actually be measured on an isl95712 regulator. t1(s) is the total loop gain of the voltage loop and the droop loop. it always has a higher crossover frequency than t2(s), therefore has a higher impact on system stability. t2(s) is the voltage loop gain with closed droop l oop, thus having a higher impact on output voltage response. v cn r sen n ------------- i o ? = (eq. 35) i droop 5 4 -- - 1 r i ----- r sen n ------------- i o ? ? ? = (eq. 36) r i 5 4 -- - r sen i o ? ni droop ? --------------------------- ? = (eq. 37) r i 5 4 -- - r sen i omax ? ni droopmax ? -------------------------------------- ? = (eq. 38) ll v droop i o ------------------ - 5 4 -- - r droop r i ------------------- ? r ntcnet r ntcnet r sum n -------------- - + ------------------------------------------ dcr n ------------- ? ? == (eq. 39) ll v droop i o ------------------ - 5 4 -- - r sen r droop ? nr i ? --------------------------------------- ? == (eq. 40) r droop i o i droop ---------------- ll ? = (eq. 41) figure 26. voltage regulator equivalent circuit i o v o vid z out (s) = ll load vr figure 27. loop gain t1(s) measurement set-up q2 q1 l i o c out v o v in gate driver comp mod. load line slope ea vid channel b channel a excitation output isolation transformer 20 loop gain = channel b channel a network analyzer + + + -
isl95712 fn8566 rev 1.00 page 30 of 35 november 2, 2015 design the compensator to get stable t1(s) and t2(s) with sufficient phase margin and an output impe dance equal to or smaller than the load line slope. current balancing refer to figures 13 through 19 for information on current balancing. the isl95712 achieves current balancing through matching the isen pin voltages. r isen and c isen form filters to remove the switching ripple of the phase node voltages. it is recommended to use a rather long r isen c isen time constant, such that the isen voltages ha ve minimal ripple and represent the dc current flowing through the inductors. recommended values are r s = 10k and c s =0.22f. thermal monitor component selection the isl95712 features two pins, ntc and ntc_nb, which are used to monitor motherboard temperature and alert the amd cpu if a thermal issues arises. the basic function of this circuitry is outlined in the ? thermal monitor [ntc, ntc_nb] ? on page 25 . figure 29 shows the basic configuration of the ntc resistor, r ntc , and offset resistor, r s , used to generate the warning and shutdown voltages at the ntc pin. as the board temperature rises, the ntc thermistor resistance decreases and the voltage at the ntc pin drops. when the voltage on the ntc pin drops below the thermal warning threshold of 0.640v, then vr_hot_l is pulled low. when the amd cpu detects vr_hot_l has gone low, it will begin throttling back load current on both outputs to reduce the board temperature. if the board temperature continues to rise, the ntc thermistor resistance will drop further and the voltage at the ntc pin could drop below the thermal shutdown threshold of 0.580v. once this threshold is reached, the isl95712 shuts down both core and northbridge vrs indicating a ther mal fault has occurred prior to the thermal fault counter triggering a fault. selection of the ntc thermistor can vary depending on how the resistor network is configured. th e equivalent resistance at the typical thermal warning threshold voltage of 0.64v is defined in equation 42 . the equivalent resistance at the typical thermal shutdown threshold voltage of 0.58v required to shutdown both outputs is defined in equation 43 . the ntc thermistor value correlates to the resistance change between the warning and shutdown thresholds and the required temperature change. if the warning level is designed to occur at a board temperature of +100c and the thermal shutdown level at a board temperature of +105c, th en the resistance change of the thermistor can be calculated. for example, a panasonic ntc thermistor with b = 4700 has a resi stance ratio of 0.03939 of its nominal value at +100c and 0.03308 of its nominal value at +105c. taking the required re sistance change between the thermal warning threshold and the shutdown threshold and dividing it by the change in resi stance ratio of the ntc thermistor at the two temperatures of interest, the required resistance of the ntc is defined in equation 44 . the closest standard thermistor to the value calculated with b = 4700 is 330k . the ntc thermistor part number is ertj0ev334j. the actual resistan ce change of this standard thermistor value between the warning threshold and the shutdown threshold is calculated in equation 45 . figure 28. loop gain t2(s) measurement set-up q2 q1 l i o c o v o v in gate driver comp mod. load line slope ea vid channel b channel a excitation output isolation transformer 20 loop gain = channel b channel a network analyzer + + + - ? ntc r ntc 30a internal to isl95712 figure 29. thermal monitor feature of the isl95712 r s monitor r +v warning 640mv shutdown 580mv vr_hot_l 330k 8.45k 0.64v 30 ? a --------------- - 21.3k ? = (eq. 42) 0.58v 30 ? a --------------- - 19.3k ? = (eq. 43) 21.3k ? 19.3k ? C ?? 0.03939 0.03308 C ?? ----------------------------------------------------- - 317k ? = (eq. 44) 330k ? 0.03939 ? ?? 330k ? 0.03308 ? ?? C 2.082k ? = (eq. 45)
isl95712 fn8566 rev 1.00 page 31 of 35 november 2, 2015 since the ntc thermistor resistan ce at +105c is less than the required resistance from equation 43 , additional resistance in series with the thermistor is required to make up the difference. a standard resistor, 1% toleranc e, added in series with the thermistor will increase the voltage seen at the ntc pin. the additional resistance requ ired is calculated in equation 46 . the closest, standard 1% tolerance resistor is 8.45k . the ntc thermistor is placed in a hot spot on the board, typically near the upper mosfet of channel 1 of the respective output. the standard resistor is placed next to the controller. layout guidelines pcb layout considerations power and signal layers placement on the pcb as a general rule, power layers should be close together, either on the top or bottom of the board, with the weak analog or logic signal layers on the opposite side of the board. the ground-plane layer should be adjacent to the signal layer to provide shielding. component placement there are two sets of critical components in a dc/dc converter; the power components and the small signal components. the power components are the most critical because they switch large amounts of energy. the small signal components connect to sensitive nodes or supply crit ical bypassing current and signal coupling. the power components should be placed first and these include mosfets, input and output capacitors, and the inductor. it is important to have a symmetrical layout for each power train, preferably with the controller located equidistant from each power train. symmetrical layout allows heat to be dissipated equally across all power trains. keeping the distance between the power train and the control ic short helps keep the gate drive traces short. these drive signals include the lgate, ugate, pgnd, phase and boot. when placing mosfets, try to keep the source of the upper mosfets and the drain of the lower mosfets as close as thermally possible (see figure 30 ). input high-frequency capacitors should be placed close to the drain of the upper mosfets and the source of the lower mosfets. place the output inductor and output capacitors between the mosfets and the load. high-frequency output decoupling capacitors (ceramic) should be placed as close as possible to the decoupling target (microprocessor), making use of the shortest connection paths to any internal planes. place the components in such a way that the area under the ic has less noise traces with high dv/dt and di/dt, such as gate signals and phase node signals. table 15 shows layout considerations for the isl95712 controller by pin. 19.3k ? 10.916k ? C 8.384k ? = (eq. 46) figure 30. typical power component placement inductor vias to ground plane vin vout phase node gnd output capacitors low-side mosfets input capacitors schottky diode high-side mosfets
isl95712 fn8566 rev 1.00 page 32 of 35 november 2, 2015 12 ntc the ntc thermistor must be placed cl ose to the thermal source that is monitored to determine core thermal throttling. placement at the hottest spot of the core vr is recommended . additional standard resistors in the resistor network on this pin should be placed near the ic. 13 isen4 each isen pin has a capacitor (c isen ) decoupling it to v sumn and then through another capacitor (c vsumn ) to gnd. place c isen capacitors as close as possible to the co ntroller and keep the following loops small: 1. any isen pin to another isen pin 2. any isen pin to gnd the red traces in the following drawing show the loops to be minimized. 14 isen3 15 isen2 16 isen1 17 isump place the current sensing circuit in general proximity of the controller. place capacitor c n very close to the controller. place the ntc thermistor next to core vr channel 1 indu ctor so it senses the inductor temperature correctly. each phase of the power stage sends a pair of v sump and v sumn signals to the controller. run these two signals traces in parallel fashion with decent width (>20mil). important: sense the inductor current by routing the sensing ci rcuit to the inductor pads. if possible, route the traces on a different layer from the inductor pad layer and use vias to conn ect the traces to the center of the pads. if no via is allowe d on the pad, consider routing the traces into the pads from th e inside of the inductor. the following drawings show the two preferred ways of routing current sensing traces. 18 isumn 19 vsen place the filter on these pins in close proximity to the controller for good coupling. 20 rtn 21 fb place the compensation components in general proximity of the controller. 22 vdd a high quality, x7r dielectric mlcc capacitor is recommended to decouple this pin to gnd. place the capacitor in close proximity to the pin with the fi lter resistor nearby the ic. 23 pgood no special consideration. 24 comp place the compensation components in general proximity of the controller. 25 boot1 use a wide trace width (>30mil). avoid routing any sensitiv e analog signal traces close to or crossing over this trace. table 15. layout considerations for the isl95712 controller (continued) pin number symbol layout guidelines v isen3 l 3 r isen isen2 isen1 l 2 l 1 r isen r isen phase1 phase2 phase3 r o r o r o gnd c isen c isen c isen c vsumn v sumn isen 4 l 3 r isen phase1 r o c isen inductor current-sensing traces vias inductor current-sensing traces
isl95712 fn8566 rev 1.00 page 33 of 35 november 2, 2015 26 phase1 these two signals should be routed together in parallel. each trace shou ld have sufficient width (>30mil). avoid routin g these signals near sensitive analog signal traces or crossing over them. routing phase1 to the core vr channel 1 high-side mosfet source pin instead of a general connection to phase1 copper is recommended for better performance. 27 ugate1 28 lgate1 use sufficient trace width (>30mil). avoid routing this signal near any sensitiv e analog signal traces or crossing over them. 29 boot2 use a wide trace width (>30mil). avoid routing any sensitiv e analog signal traces close to or crossing over this trace. 30 phase2 these two signals should be routed together in parallel. each trace shou ld have sufficient width (>30mil). avoid routin g these signals near sensitive analog signal traces or crossing over them. routing phase2 to the core vr channel 2 high-side mosfet source pin instead of a general connection to phase2 copper is recommended for better performance. 31 ugate2 32 vddp a high quality, x7r dielectric mlcc capacitor is recommended to decouple this pin to gnd. place the capacitor in close proximity to the pin. 33 lgate2 use sufficient trace width (>30mil). avoid routing this signal near any sensitiv e analog signal traces or crossing over them. 34 lgate1_nb use sufficient trace width (>30mil ). avoid routing this signal near any sens itive analog signal traces or crossing o ver them. 35 phase1_nb these two signals should be rout ed together in parallel. each trace should have sufficient width (>30mil). avoid rou ting these signals near sensitive analog sign al traces or crossing over them. rout ing phase1_nb to the high-side mosfet source pin instead of a general connection to the ph ase1_nb copper is recommended for better performance. 36 ugate1_nb 37 boot1_nb use a wide trace width (>30mil). avoid routing any sensit ive analog signal traces close to or crossing over this trac e. 38 pwm3 no special considerations. 39 pwm4 no special considerations. 40 pwm2_nb no special considerations. 41 pwm3_nb no special considerations. 42 i2clk use good signal integrity practices 43 i2data use good signal integrity practices 44 prog no special considerations. 45 pgood_nb no special consideration. 46 comp_nb place the compensation components in general proximity of the controller. 47 fb_nb 48 vsen_nb place the filter on this pin in close proximity to the controller for good coupling. 49 isumn_nb place the current sensing circuit in general proximity of the controller. place capacitor c n very close to the controller. place the ntc thermistor next to nb vr channel 1 indu ctor so it senses the inductor temperature correctly. each phase of the power stage sends a pair of v sump and v sumn signals to the controller. run these two signals traces in parallel fashion with decent width (>20mil). important: sense the inductor current by routing the sensing ci rcuit to the inductor pads. if possible, route the traces on a different layer from the inductor pad layer and use vias to conn ect the traces to the center of the pads. if no via is allowe d on the pad, consider routing the traces into the pads from th e inside of the inductor. the following drawings show the two preferred ways of routing current sensing traces. 50 isump_nb 51 isen1_nb each isen pin has a capacitor (c isen ) decoupling it to vsumn_nb, then through another capacitor (c vsumn_nb ) to gnd. place c isen capacitors as close as possible to the co ntroller and keep the following loops small: 1. any isenx_nb pin to another isenx_nb pin 2. any isenx_nb pin to gnd 52 isen2_nb table 15. layout considerations for the isl95712 controller (continued) pin number symbol layout guidelines inductor current-sensing traces vias inductor current-sensing traces
fn8566 rev 1.00 page 34 of 35 november 2, 2015 isl95712 intersil products are manufactured, assembled and tested utilizing iso9001 quality systems as noted in the quality certifications found at www.intersil.com/en/suppor t/qualandreliability.html intersil products are sold by description on ly. intersil may modify the circuit design an d/or specifications of products at any time without notice, provided that such modification does not, in intersil's sole judgment, affect the form, fit or function of the product. accordingly, the reader is cautioned to verify that datasheets are current before placing orders. information fu rnished by intersil is believed to be accu rate and reliable. however, no responsib ility is assumed by intersil or its subsidiaries for its use; nor for any infrin gements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of intersil or its subsidiaries. for information regarding intersil corporation and its products, see www.intersil.com for additional products, see www.intersil.com/en/products.html ? copyright intersil americas llc 2014-2015. all rights reserved. all trademarks and registered trademarks are the property of their respective owners. about intersil intersil corporation is a leading provider of innovative power ma nagement and precision analog so lutions. the company's product s address some of the largest markets within the industrial and infrastr ucture, mobile computing and high-end consumer markets. for the most updated datasheet, application notes, related documentatio n and related parts, please see the respective product information page found at www.intersil.com . you may report errors or suggestions for improving this datasheet by visiting www.intersil.com/ask . reliability reports are also av ailable from our website at www.intersil.com/support . revision history the revision history provided is for informational purposes only and is believed to be accurate, but not warranted. please go t o web to make sure you have the latest revision. date revision change november 2, 2015 fn8566.1 on page 1 under features, added ?serial vi d clock frequency range 100khz to 25mhz? below ?supports amd s vi 2.0 serial data bus interface and pmbus?. updated package outline drawing l52.6x6a to the latest revision. changes are as follows: -added tolerance values. march 26, 2014 fn8566.0 initial release
isl95712 fn8566 rev 1.00 page 35 of 35 november 2, 2015 package outline drawing l52.6x6a 52 lead quad flat no-lead plastic package chamfered corner leads rev 1, 7/14 bottom view detail "x" side view typical recommended land pattern top view located within the zone indicated. the pin #1 identifier may be unless otherwise specified, tolerance: decimal 0.05 tiebar shown (if present) is a non-functional feature. the configuration of the pin #1 identifier is optional, but mus t be between 0.15mm and 0.30mm from the terminal tip. dimension applies to the metallized terminal and is measured dimensions in ( ) for reference only. dimensioning and tolerancing conform to asme y14.5m-1994. 6. either a mold or mark feature. 3. 5. 4. 2. dimensions are in millimeters. 1. notes: (48x0.40) (52x0.20) (5.80 typ) ( 4.70) (52x0.60) index area (4x) 0.15 pin 1 6 a b c 0.900 0.10 0.2 ref 0.05 max. 0.00 min. 5 seating plane 0.10 c 0.08 c c 52x0.40 27 26 14 39 40 0.40 48x 4x 4.8 52 13 see detail "x" 52x0.20 4 4.70 0.10 pin #1 6 1 0.165 typ. r0.100 typ. 0.165 typ 4x see index area 0.10 c a b 0.05 c detail "y" detail "y" 6.00 0.05 6 . 0 0 0 . 0 5

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